Methods, apparatus, and systems for coding with constrained interleaving

ABSTRACT

Serially-concatenated codes are formed in accordance with the present invention using a constrained interleaver. The constrained interleaver cause the minimum distance of the serial concatenated code to increase above the minimum distance of the inner code alone by adding a constraint that forces some or all of the distance of the outer code onto the serially-concatenated code. This allows the serially-concatenated code to be jointly optimized in terms of both minimum distance and error coefficient to provide significant performance advantages. Constrained interleaving can be summarized in that it: 1) uses an outer code that is a block code or a non-recursive convolutional code, and as such, there are multiple codewords present in the constrained interleaver, 2) selects a desired MHD, 3) selects an interleaver size and a set of predefined interleaver constraints to prevent undesired (low-distance) error events so as to achieve the desired MHD, and 4) performs uniform interleaving among the allowable (non-constrained) positions, to thereby maximize or otherwise improve the interleaver gain subject to the constraints imposed to maintain the desired MHD.

This patent application is a continuation of co-pending U.S. patent application Ser. No. 13/694,014, filed Oct. 22, 2012 which is a continuation-in-part of U.S. patent application Ser. 12/926,539, filed Nov. 24, 2010 which claims priority to U.S. provisional patent application No. 61/344,675, entitled “Encoding and decoding using constrained interleaving,” filed Sep. 10, 2010.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to communication encoders, decoders, transmitters, receivers, and systems. More particularly, aspects of the invention relate a family of encoders and a family of decoders that make use of constrained interleaving with various forms of serially concatenated codes.

2. Description of the Related Art

Various forms of concatenated codes are known in the art. Turbo codes are widely used and represent parallel concatenated codes. In the literature, interleaver design has been discussed with both parallel and serial concatenation but mainly in connection with parallel concatenated codes. For example, see [1] J. Yu, M.-L. Boucheret, R. Vallet, A. Duverdier and G. Mesnager, “Interleaver design for serial concatenated convolutional codes”, IEEE Commun. Letters, Vol. 8, No. 8, pp. 523-525, August 2004; [2] F. Daneshgaran, M. Laddomada and M. Mindin, “Interleaver design for serially concatenated convolutional codes: Theory and application”, IEEE Trans. On Inform Theory, vol. 50, No. 6, pp. 1177-1188, June 2004; [3] H. R. Sadjadpour, N. J. A. Sloane, G. Nebe and M. Salehi, “Interleaver design for turbo codes”, in proc. ISIT, pp. 453, June 2000; and [4] H. R. Sadjadpour, N.J.A. Sloane, M. Salehi and G. Nebe, “Interleaver design for turbo codes”, IEEE Journal of selected areas in Commun., vol. 19, pp. 831-837, May 2001, as supplied on the IDS herewith.

The above design approaches start from uniform interleaving and modify the uniform interleaver's probability distribution according to various rules. A “uniform Interleaver” is defined by a randomization operation of the form Output=Rand(Input) where Input and Output represent respective vectors of N elements, and Rand is a function that pseudo randomly permutes the order of the elements in the vector Input. In most cases, the prior art uniform interleaver is a “bit interleaver” and the elements of Input and Output represent bits. That is, the uniform interleaver is used to randomize the order of a set of input bits to create a randomized-ordered set of output bits.

While uniform interleaving or its variants as referenced above may be the best ways to construct interleavers for use with parallel concatenated codes, it would be desirable to have a different form of interleaving that takes advantage of correlations that exist in coded bits that have been formed via serial concatenation encoding. Unlike parallel concatenation, where interleaving is performed on pure uncorrelated information bits which are usually independent, in the case of serial concatenation, the interleaver is used on the coded bits of the outer code which are correlated due to the outer code. It would be desirable to have an interleaving technique for use with serial concatenation that exploits the correlation of the coded bits introduced by the outer code. It would be desirable if interleavers designed to exploit that correlation were able to have much shorter interleaver lengths while being able to perform as well or better than much longer interleavers that are designed using prior art approaches that amount to variants of uniform interleaving.

FIG. 1 shows a prior art turbo encoder. As can be seen from the parallel structure of FIG. 1, Turbo encoders are based on parallel concatenation. The message bits are replicated and processed on three (or in general, more) paths. The first path has no coding, the second path encodes the message bits with Encoder #1 which is usually a convolutional code, and the third path uniform interleaves the message bits and then encodes the interleaved message bits with Encoder #2. Three times as many bits are produced using this parallel approach, resulting in a rate 1/3 code. Code puncturing can be optionally used to increase the rate of the concatenated code. Turbo codes are usually decoded using an iterative decoder structure similar to the one shown in FIG. 5 with the constrained interleavers/deinterleavers replaced with uniform interleavers/deinterleavers. The soft decoder of FIG. 5 uses the well known BJCR algorithm or some other type of soft decoding algorithm in its soft decoding blocks.

FIG. 2 shows the serial structure of a prior art serial concatenated encoder. The message bits are first encoded by an outer encoder, then the outer-encoded bits are sent through a uniform interleaver, and the interleaved outer-encoded bits are next passed through an inner encoder. In general, more than two component codes can be concatenated together, but, without loss of generality, the discussion herein focuses on embodiments that make use of two serially concatenated component codes. The concepts presented herein can be extrapolated to these higher order cases by induction. In many practical cases the outer code is a block code or a nonrecursive convolutional code and the inner code is a recursive convolutional code. Serial concatenated codes are also usually decoded using an iterative decoder structure similar to the one shown in FIG. 5 with the constrained interleavers/deinterleavers replaced with uniform interleavers/deinterleavers.

It is known that serially concatenated codes and parallel concatenated codes can both be designed to achieve interleaver gain. “Interleaver gain” is defined as a reduction in the bit error rate as the interleaver length, N, is increased. This occurs because certain dominant error coefficients in the probability of error expression are reduced as N is increased. It is known in the art that serially concatenated codes can be designed to perform better than parallel concatenated codes with similar parameters. Serial concatenation can employ component codes that are block and/or convolutional codes. General design rules of serially concatenated codes are well known. It is generally advantageous to use an outer code that has a high minimum Hamming distance and to employ a recursive inner code. However, it is also known that even though the traditional method of serial concatenation is done using recursive inner codes, block codes can also be effectively used for the inner code as well; for example, see [5] M. Sikora and J. Costello, Jr., “Serial concatenation with simple block inner codes”, in proc. ISIT, pp. 1803-1807, July 2006. Serially concatenated codes can be decoded using iterative soft decoding of inner and outer codes (using a structure similar to that shown in FIG. 5 but with a uniform interleaver such as one that may be implemented using a randomization function with a uniform distribution or e.g., see [1]-[4]).

More background information on serial concatenated codes that supports the discussion in the above paragraph can be found in: [6] S. Benedetto, D. Divsalar, G. Montorsi and F. Pollara, “Serial concatenation of interleaved codes: Performance analysis, design and iterative decoding”, IEEE Trans., Inform. Theory, vol. 44, pp. 909-926, May 1998 [7] S. Benedetto, D. Divsalar, G. Montrosi and F. Pollara, “Analysis, design, and iterative decoding of double serially concatenated codes with interleavers”, IEEE Journal on Selected Areas in Commun., vol. 16, No. 2, pp. 231-244. February 1998; [8] S. Benedetto and G. Montrosi, “Iterative decoding of serially concatenated convolutional codes”, Electronics Letters, vol. 32, No. 13, pp. 1186-1188, June 1996; [9] S. Benedetto, D. Divsalar, G. Montrosi and F. Pollara, “A soft-input soft-output APP module for iterative decoding of concatenated codes”, IEEE Commun. Letters, pp. 22-24. January 1997; and [10] J. Hagenauer, E. Offer, and L. Papke, “Iterative decoding of binary block and convolutional codes”, IEEE Trans. Inform. Theory, vol. 42, pp. 429-445, March 1996; all of which are included on the IDS submitted herewith. For further background information on both serial and parallel concatenated codes, also see S. Lin and D. Costello, Jr., Error Control Coding: Fundamentals and Applications, 2^(nd) Ed., Pearson Prentice-Hall, 2004.

Multi-dimensional SPC codes are also well known in the art. It is known that uniform interleaving can be applied in some cases to improve the performance of these codes, but not in other cases such as 2-dimensional SPC codes. See for example: [11] D. M. Rankin and T. A. Gulliver, “Single parity check product codes”, IEEE Trans. On Commun., vol. 49, pp. 1354-1362, August 2001; [12] X. R. Ma and Y. Y. Xu, “Iterative decoding of parallel and serial concatenated single parity check product codes”, Electronics Letters, vol. 42, No. 15, pp. 869-870, July 2006; [13] L. Ping, S. Chan and K. L. Yeung, “Efficient soft-in-soft-out sub-optimal decoding rule for single parity check codes”, Electronics Letters, vol. 33, No. 19, pp. 1614-1616, September 1997; [14] D. Rankin and A. Gulliver, “Randomly interleaved SPC product codes”, in Proc. ISIT, pp. 88, 2000; and [15] D. M. Rankin, T. A. Gulliver and D. P. Taylor, “Parallel and serial concatenated single parity check product codes”, EURASIP Journal on Applied Signal Processing, pp. 775-783, January 2005.

It would be desirable to improve the performance of 2-dimensional SPC codes using an inventive constrained interleaver. It would be desirable to be able to use shorter interleavers to provide the same or improved performance over SPCs that currently use uniform interleavers.

In both parallel and serially concatenated codes, the design objective traditionally has been to focus on the interleaver gain that affects the error coefficient as opposed to the minimum distance of the resulting concatenated code. This makes sense when it is tolerable to employ long interleavers. However, due to the increase in the complexity, memory requirements and delay caused by long interleavers, in practice it is preferable to avoid making the size of the interleaver too large. While it would be desirable to consider the minimum distance of the overall code for short to moderate interleaver sizes, it is conventional wisdom that the joint consideration of both the minimum distance and the interleaver design is too difficult to handle [6]. It would be desirable to develop a technique to improve bit error rate performance by introducing an interleaver constraint that has the effect of jointly optimizing or otherwise jointly considering both the minimum distance and the reduction of dominant error coefficients. It would be desirable to have a new technology that used such interleaver constraints to design more efficient encoders and decoders for various forms of serially concatenated codes.

SUMMARY OF THE INVENTION

The present invention provides a family of encoders, decoders, transmitters, receivers, and methods, apparatus and systems employing the same. Aspects of the present invention subject an interleaver to a selected constraint. The constraint is selected to cause a measure of minimum distance in a serial concatenated code to increase above that of the same serially concatenated code if uniform interleaving were used instead of the constrained interleaving. The net effect of a constrained interleaver is to improve the bit error rate performance over traditionally interleaved serial concatenated codes at a given interleaver length. This allows much shorter interleavers to be used and allows new types of serial concatenated codes to be constructed that would not have their performance benefits if prior art uniform interleaving were applied.

Constrained interleaving can be summarized in that it: 1) uses an outer code that is a block code or a non-recursive convolutional code, and as such, there are multiple codewords present in the constrained interleaver, 2) selects a desired MHD, 3) selects an interleaver size and a set of predefined interleaver constraints to prevent undesired (low-distance) error events so as to achieve the desired MHD, and 4) performs uniform interleaving among the allowable (non-constrained) positions, to thereby maximize or otherwise improve the interleaver gain subject to the constraints imposed to maintain the desired MHD.

A first aspect of the present invention relates to encoder and transmitter apparatus, methods and systems. An outer encoder is configured to transform a sequence of input bits to a sequence of outer encoded bits. The sequence of outer-encoded bits is encoded in accordance with an outer code. A constrained interleaver is configured to implement a permutation function to permute the order of the outer-encoded bits to produce a constrained-interleaved sequence of outer-encoded bits. An inner encoder is configured to transform the constrained-interleaved sequence of outer-encoded bits to a sequence of inner-encoded bits. The sequence of inner-encoded bits is encoded in accordance with an inner code. The sequence of inner-encoded bits constitutes a serially-concatenated sequence of bits that incorporates coding from both the inner code and the outer code in accordance with a serially-concatenated code that has a minimum distance of d_(sc), the outer code has a minimum distance of d_(o) and the inner code has a minimum distance of d_(i). The constrained interleaver's permutation function implements a constraint in order to enforce d_(i)<d_(sc)≦d₀d_(i). The distances d_(sc), d₀ and d_(i) can be representative of Hamming distances. In some embodiments, Euclidian distances can also be considered. While some prior art approaches may have achieved distances in the range of d_(i)<d_(sc)≦d₀d_(i), d_(sc) would have been much closer to d_(i) than d₀d_(i), and this would have been due to properties of the component codes as opposed to a property of the interleaver or any constraint met by the interleaver.

In transmitter embodiments, a signal mapper is also provided that is configured to map the sequence of inner-encoded bits to a transmission signal. The signal mapper can be selected such that a measure of Euclidian distance in the serially concatenated code is greater than a corresponding measure of Euclidian distance of the serially concatenated code when implemented with a uniform interleaver.

As discussed in further detail herein, the constraint implemented by the constrained interleaver is chosen to preserve a distance provided by the outer code. The advantage of this distance is generally destroyed by a uniform interleaver, i.e., the distance of the outer code does not improve the distance of the prior art serially concatenated codes which is usually d_(sc)=d_(i). In many embodiments of the present invention, the interleaver constraint is selected to enforce d_(sc)=d₀d_(i). In some alternative embodiments, the permutation function implemented by the constrained interleaver is constrained to enforce a minimum distance d_(sc)<d₀d_(i) such that a measure of bit error probability at least one specified signal to noise ratio is less than the measure of bit error probability at the at least one specified signal to noise ratio for a second constraint that enforces d_(sc)=d₀d_(i). The measure of bit error probability at the at least one specified signal to noise ratio is a function of at least one error coefficient and the reduction of the measure of the bit error probability using the constraint is caused by a reduction in an effect of the at least one error coefficient.

Constrained interleaving can be used in serially concatenated codes of various types, for example, where the outer code is a block code or a non-recursive convolutional code, or where the inner code is a non-recursive convolutional code or where the inner code is a recursive convolutional code. The permutation function of the constrained interleaver can be implemented efficiently at runtime using a stored vector of pointers in accordance with table lookup processing. That is, the reordering operation of the constrained interleaver (and/or constrained deinterleaver) is implemented by incrementing through the pointer array which encodes the reordering rule of the constrained interleaver or deinterleaver.

As is discussed herein, especially when inner recursive convolutional codes are used, additional constraints can be added to force d_(sc)>d₀d_(i). However, these additional constraints lower the total number of allowable interleaver combinations and lower the interleaver gain. For a given application to include a particular set of codes and a particular signal mapping policy, numerical simulations can be used to determine if the additional constraints would improve the overall coding gain. Several worked out examples and families of embodiments are detailed hereinbelow for the case where d_(sc)>d₀d_(i).

Herein, a constrained interleaver that is designed to enforce d_(sc)=d₀d_(i) is referred to as a “constrained interleaver type 1” or “CI-1.” For the case of SCCC's (the inner code is a recursive convolutional code) additional constraints can be added to enforce the MHD of the concatenation to be increased beyond d_(sc)=d₀d_(i) so that d_(sc)>d₀d_(i). Herein, a constrained interleaver that is designed to enforce d_(sc)>d₀d_(i) is referred to as a “constrained interleaver type 2” or “CI-2.” A constrained interleaver that trades off distance for interleaver gain to achieve d_(i)<d_(sc)<d₀d_(i) is referred to as a “constrained interleaver type 0” or “CI-0.”

Constrained interleaving can also be applied to parallel concatenation (such as turbo codes). However, this can only guarantee that the second constituent code can spread the error events. As a result, it cannot guarantee the product of the distances for the concatenation. However, due to the improvement in the second constituent code, the constrained interleaving methods, apparatus, and systems presented herein can improve performance of parallel concatenated codes over uniform interleaving. In the case of the parallel concatenated codes, the additional constraints described in the above paragraph and in later in the description of the preferred embodiments can also be used. This provides a means to improve interleavers such as those disclosed in U.S. Pat. No. 6,857,087 due to a higher interleaver gain and due to having a target overall minimum distance to control the design.

Another aspect of the present invention involves a receiver and decoder methods, apparatus, and systems. In this patent application, the term “function instantiation” should be given a particular meaning. “Function instantiation” means an embodiment of a function implemented in hardware or software. In the case of software, a given function may be written as a piece of software, but this piece of software might be called many times using different sets of input parameters. Each call to the single function would involve a “function instantiation.” In hardware, a given module that implements a function and is passed input parameters to implement the function differently can have multiple function instantiations even though only one hardware functional unit can be located in a given device.

An aspect of the present invention involves a decoder or receiver that decodes a serial concatenated code formed via constrained interleaving similar to the one discussed above, i.e., where the outer code has a minimum distance of d₀, the inner code has a minimum distance of d_(i), and the permutation function implemented by a constrained interleaver function instantiation is constrained to preserve a distance provided by the outer code to enforce d_(i)<d_(sc)≦d₀d_(i).

In various exemplary receiver apparatus and method embodiments, a signal conditioning unit is coupled to receive a received signal and operative to produce therefrom a vector of bit metrics. The received signal is a received version of a transmitted signal that was serially-concatenated encoded by a serially-concatenated encoder that coupled an outer encoded bit stream via a first constrained interleaver to an inner encoder.

A first soft decoder function instantiation is provided that is operative to soft decode its input to generate a vector of extrinsic information. The first soft decoder function instantiation decodes in accordance with the inner code and the input is initially the vector of bit metrics and subsequently an interleaved vector of inner-code soft-decoded extrinsic information. A first constrained deinterleaver function instantiation is operative to deinterleave the vector of bit metrics in accordance with an inverse permutation function that is the inverse of a permutation function employed by the first constrained interleaver. The first constrained deinterleaver function instantiation produces a deinterleaved vector of bit metrics. A second constrained deinterleaver function instantiation is provided that is operative to deinterleave the vector of inner-code soft-decoded extrinsic information in accordance with the inverse permutation function. The second constrained deinterleaver function instantiation produces a deinterleaved vector of inner-code soft-decoded extrinsic information. A second soft decoder function instantiation is operative to soft decode the deinterleaved vector of inner-code soft-decoded extrinsic information using the deinterleaved vector Of bit metrics to generate a vector of outer-code soft-decoded extrinsic information. The second soft decoder function instantiation decodes in accordance with the outer code. A stopping criterion function instantiation is operative to determine whether a measure of the outer-code soft-decoded extrinsic information has successfully passed a convergence test. Preferably when the convergence test fails, a constrained interleaver function instantiation is operative to interleave the vector of outer-code soft-decoded extrinsic information in accordance with the permutation function. The interleaver function instantiation produces an interleaved vector of outer-code soft-decoded extrinsic information. The inventive decoder or receiver method or apparatus applies iterative decoding to iteratively apply the first and second soft decoders until the convergence test has been met, and once the convergence test has been met, to then provide a decoded bit sequence produced by the second soft decoder function instantiation.

A second class of decoder or receiver embodiments decode the same kind of signal as discussed in the receiver/decoder embodiment discussed above. In this second class of embodiments, a list Viterbi decoder function instantiation is provided that is operative to provide a p^(th) decoded sequence estimate. The list Viterbi decoder decodes in accordance with a list Viterbi algorithm based upon the inner code and p is a positive integer that is incremented as p=1, 2, . . . MaxList, where MaxList≧1 is a predefined maximum number of sequence estimates that will be output from the list Viterbi decoder. An outer code match detector function instantiation is provided that is operative to determine whether the p^(th) decoded sequence estimate has successfully passed a convergence test that is based on a measure of the outer code. The receiver apparatus couples as an output the first decoded sequence estimate sequence that successfully passes a convergence test. One preferred embodiment uses parallel processing to generate the different Viterbi list sequences in parallel. In a preferred embodiment for sequential based processor embodiments, the list Viterbi decoder sequentially outputs one of the decoded sequence estimates at a time and stops decoding as soon as the match detector indicates that the convergence test has been satisfied at some value of p<MaxList.

BRIEF DESCRIPTION OF THE FIGURES

The various novel features of the present invention are illustrated in the figures listed below and described in the detailed description that follows.

FIG. 1 illustrates a prior art turbo encoder that generates a parallel concatenated code.

FIG. 2 illustrates a prior art encoder that generates a serially concatenated code.

FIG. 3 is a block diagram of an embodiment of an encoder that generates a serially concatenated block code (SC-BC) using a constrained interleaver and block encoders to implement both the inner and outer codes.

FIG. 4 is a block diagram of an embodiment of an encoder that generates a serially concatenated code with an inner recursive convolutional code (SC-IRCC) using a constrained interleaver and a recursive convolutional code as the inner code.

FIG. 5 is a block diagram of an embodiment of a soft iterative decoder that makes use of constrained interleaving and constrained deinterleaving to decode SC-BCs or SC-IRCCs that have been generated in accordance with constrained interleaving.

FIG. 6 is a block diagram of an embodiment of list Viterbi decoder based decoder system used to decode a serially concatenated code as produced by one of the encoders of FIG. 3 or FIG. 4 or their variants or equivalents.

FIG. 7 is a block diagram of an exemplary communication system and method including two transmitters and two receivers that make use of the serial concatenation coding with constrained interleaving in order to communicate between communication endpoint stations.

FIG. 8 is a flow chart that illustrates the operation of a constrained interleaver for operation with the block-code based encoder of FIG. 3.

FIG. 9 illustrates a constrained interleaving example using the flow chart of FIG. 6, where q=3 and m=6.

FIG. 10 shows three bit error rate performance curves that illustrate how constrained interleaving can reach the performance bound of uniform interleaving, but with a much shorter interleaver (e.g., N=16, N=40 and N=400) in an example involving serially concatenated block codes.

FIG. 11 shows bit error rate performance curves and error rate bounds of 2-D SPC codes with uniform interleaving and constrained interleaving along with that of without interleaving when the interleaver length is N=12 and N=90 in an example involving serially concatenated block codes.

FIG. 12 illustrates the r×ρn constrained interleaver array structure of a constrained interleaver designed to operate in accordance with the flow chart of FIGS. 4 and 13.

FIG. 13 is a flow chart that illustrates the operation of a constrained interleaver for operation with the inner-recursive-code based encoder of FIG. 4.

FIG. 14 shows bit error rate performance curves of uniform and constrained interleaving of an outer (7,4) Hamming code and a rate 1/2 inner recursive convolutional code when r=8 and with interleaver lengths of N=112 and N=336 and N=1008.

FIG. 15 shows bit error rate performance curves of constrained interleaving with a (7,6) outer SPC code and along with a rate 1/2 inner convolutional code and compares this to the same code implemented with uniform interleaving with interleaver lengths of N=112 and N=336 and N=1008.

FIG. 16 shows the bit error rate performance curve of a serial concatenation of two block codes, an outer (10,9) SPC and an inner (64,45) extended BCH code; the Shannon limit is also plotted.

FIG. 17 shows the bit error rate performance curve of a serial concatenation of an outer (15,10) extended Hamming code and an inner code that is a rate 2/3 punctured recursive convolutional code with 4 states; the Shannon limit is also plotted.

FIG. 18 is a block diagram of an exemplary double SCCC encoder.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In this patent application, when two block codes are serially concatenated, this is referred to as SC-BC (serially concatenated block codes). When the inner code is a convolutional code, this is referred to as SCCC (serial concatenation with a convolutional code). The acronym IRCC stands for a inner recursive convolutional code. An IRCC is a recursive convolutional code that is used as an inner code in a concatenated encoder. When the inner code is specifically a recursive convolutional code (RCC), i.e., when an IRCC is used in a serial concatenated code, this is referred to SC-IRCC (serial concatenation with an inner recursive convolutional code). As it turns out, the constrained interleavers designed for these various types of serially concatenated codes preferably require different sets of constraints to achieve the best performance. Hence the encoder and decoder designs for these various code types are described separately. Also, the term SC-CI is used to describe any of the above concatenated codes when a constrained interleaver is employed as shown in FIG. 3 and FIG. 4.

FIG. 3 illustrates an embodiment of a serial concatenated encoder and transmitter designed in accordance with the present invention. In general, more than two component codes can be concatenated together, but, without loss of generality, the discussion herein focuses on embodiments than make use of two serially concatenated component codes. The concepts presented herein can be extrapolated to these higher order cases by induction. FIG. 3 is an embodiment that makes use of an outer encoder 305 and an inner encoder 315 which are both block encoders. The message bit stream at the input can be considered to be a sequence of k-bit blocks which are each processed first by the outer encoder 305. The outer encoder 305 encodes according to a systematic (q, k) outer code with minimum distance d₀. The outer-encoded bits can be viewed as a sequence of q-bit codewords which are fed into a constrained interleaver 310. The operation of the inventive constrained interleaver 310 is discussed hereinbelow in further detail in connection with FIG. 8. The output bit stream of the constrained interleaver 310 is fed to the inner coder 315 which implements an (n, q) inner code with minimum distance d_(i). The inner coder 315 preferably implements a systematic code. Even though systematic codes are considered here by way of example, the constrained interleaving technique presented here will work equally well with non-systematic component block codes too.

The constrained interleaver 310 can be viewed a permutation function that is applied to a vector of bits to produce an output vector of bits whose order has been altered relative to the input vector in accordance with the permutation function. Constrained interleaving differs from uniform interleaving because the permutation function is selected to meet a set of constraints that are designed to jointly improve or optimize the minimum distance and dominant error coefficients of the serially concatenated code that is output from the block 315 (or 415 as discussed below). As is discussed in connection with FIG. 8, the constrained interleaver 315 may be designed or implemented using a data structure that not only includes this bit vector, but also includes a set of memory pointers that allow hardware or software to treat the bit vector as a rectangular array. In a preferred embodiment, this rectangular array is of size (q×m) where the array elements correspond to outer-encoded bits. This allows the constrained interleaver to interleave m codewords of the outer code, or equivalently mq outer coded bits. The output of the constrained interleaver is coupled to the inner encoder 315 which applies a (n, q) block code. That is, the constrained-interleaved bits are fed into the inner encoder 315 and the output of the inner encoder 315 is a codeword of the (mn, mk) serially concatenated (block) code. In other words, there are mk input bits each serially concatenated coded frame and there are M_(SC-BC)=mn serially concatenated output bits each frame, where, as previously discussed, the subscript SC-BC stands for “serial concatenated block code.” The constrained-interleaved SC-BC is generated as per FIG. 3 or its variants or equivalents.

The portion of the transmitter 300 minus the mapper 320 constitutes an encoder embodiment 300 that can be implemented independently of a mapper 320. In a transmitter embodiment, the bits of the (mn, mk) serially concatenated code are additionally coupled to the mapper 320. The mapper 320 maps the encoded bits onto a signal constellation selected for the specific embodiment. For example, the mapper can generate a binary phase shift keyed (BPSK) signal, a quadrature phase-shift keyed (QPSK) signal (either of which can be further subjected to a spreading signal in spread spectrum embodiments), a quadrature amplitude modulated (QAM) signal, a modulated optical signal, a magnetic recording channel signal, an orthogonal frequency division multiplexed (OFDM) signal or the like to be transmitted via wire, fiber optic, or wireless means. The output of the mapper is the transmitted signal, and thus the mapper 320 may generally also include frequency up-shifting, amplification, antenna and other components needed to transmit the mapped signal to a remote station, for example, as discussed in connection with FIG. 7.

It is noted that in OFDM embodiments, the mapper 320 can be a mapper that maps the concatenated encoded signal to a plurality of carriers. Alternatively, a separate transmitter 300 can be implemented for each subcarrier or for subsets of subcarriers. In such cases the mapper 320 may be implemented as a sub-portion of a larger mapper such as a fast Fourier transform unit that collects coded bits from a plurality of encoders like the encoder 300 and maps them in bulk onto a set of carriers.

In certain types of embodiments, the mapper 320 is an 8PSK mapper, a QAM mapper, a multidimensional code mapper are used in multidimensional trellis coded modulation applications, or any other kind of mapper used in trellis coded modulation. It is known that non-recursive convolutional codes and block codes behave in a similar manner in serial concatenation. That is, the inner coder 315 can be implemented as a finite-length, non-recursive, trellis encoder. In such applications, the performance of the serially concatenated code with inner non-recursive code will have a performance lower bound similar to serial concatenated codes based on block codes (as discussed in further detail below). Hence an inventive concept is to improve upon trellis coded modulation schemes by replacing the trellis code with the encoder of FIG. 3 where inner coder is implemented as a non-recursive trellis encoder and the outer code is then used to improve the performance of the trellis encoded modulation scheme. The constrained interleaver 310 will be relatively short. This modified trellis coded modulated signal can be sent over a channel or mapped onto one or more subcarriers in an OFDM embodiment.

Since constrained interleaving in some way controls the merging event, the best mapping policy for the mapper 320 constrained interleaving can differ with that of the same system 300 if the constrained interleaver 310 is implemented as uniform interleaver (i.e., no constrained need be satisfied as discussed below). The best mapping policy will depend on the component codes and the operating error rates. For example, if the code is used at very low error rates, the mapper should be selected to maximize the minimum Euclidean distance. However, if the application targets moderate error rates, then different terms other than the minimum distance terms may dominate the error rates. Hence, the mapping policy can be different from the one that generates the maximum minimum distance. A numerical search can be performed to find a mapping rule for the mapper 320 that minimizes the error rate for the application based on the operating conditions and parameters where the system will operate.

FIG. 4 illustrates a second type of embodiment of a serial concatenated encoder and transmitter designed in accordance with the present invention. In general, more than two component codes can be concatenated together, but, without loss of generality, the discussion herein focuses on embodiments that make use of two serially concatenated component codes. The concepts presented herein can be extrapolated to these higher order cases by induction. A characterizing feature of the embodiment of FIG. 4 is that it makes use of an inner encoder 415 that encodes its input bit stream in accordance with an IRCC as shown in FIG. 4. The embodiment of FIG. 4 makes use of an outer encoder 405 that encodes k-bit blocks of the message bits according to an (n, k) block code. As is discussed below, alternative embodiments can be formed where the outer encoder 405 is implemented as a non-recursive convolutional encoder. In still other embodiments the outer encoder 405 can implement a recursive convolutional code. In general, any kind of code can be used by the outer encoder 405, but block codes and non-recursive convolutional codes are believed to be the preferred embodiments at this time. However, in a broader family of embodiments of which FIG. 4 is one example, the inner encoder 415 is always implemented as an IRCC. If block 415 of FIG. 4 is altered in a way that the IRCC is replaced with a non-recursive convolutional code, this is referred to as an SCCC and such embodiments are also contemplated and discussed below.

Focusing specifically on the embodiment of FIG. 4, the message bit stream at the input can be considered to be a sequence of k-bit blocks which are each processed first by the outer encoder 405 which encodes according to a systematic (n, k) outer code with minimum distance d₀ ^(f). The outer-encoded bits can be viewed as a sequence of n-bit codewords which are fed into a constrained interleaver 410. The operation of the inventive constrained interleaver 410 is discussed hereinbelow in further detail in connection with FIG. 13. As discussed in more detail below, the implementation of the constrained interleaver 410 is different than the implementation of the constrained interleaver 310. That is, the constrained interleavers 310 and 410 implement different sets of constraints in order to improve bit error rate performance in the presence of the different types of inner codes implemented by the inner encoders 315 and 415. The constraints are designed to jointly increase the concatenated code's minimum distance and to reduce the effect of dominant error coefficients. The bit error rate performance is a function of both the minimum distance of the concatenated code and the error coefficients as is discussed in further detail below.

The output bit stream of the constrained interleaver 410 is fed to the inner coder 415 which implements the IRCC with a minimum distance d_(f) ^(i). The constrained interleaver 410 can also be viewed as a permutation function that operates on a vector of bits, but this time the length of the vector is rρn where n is defined as above, r corresponds to the number or rows in the constrained interleaver 415, and ρ corresponds to the number of codewords of the outer code per row in the constrained interleaver 415. Conceptually, the bit vector that the constrained interleaver 415 permutes can be viewed as rectangular array is of size (r×ρn) where the array elements correspond to outer-encoded bits, loaded into the array in row-major order. Equivalently, this rectangular array can be viewed as an array of size (r×ρ) where the array elements correspond to n-bit codewords. This allows the constrained interleaver to interleave rρn outer coded bits. The output of the constrained interleaver is coupled to the inner encoder 415 which encodes according to the IRCC. That is, the constrained-interleaved bits are fed into the inner encoder 415 and the output of the inner encoder 415 is a valid coded sequence of the SC-IRCC, i.e., a constrained-interleaved serially concatenated code that employs an IRCC.

As is discussed in connection with FIG. 13, the constrained interleaver 415 may be implemented as a data structure that not only includes this bit vector, but also includes a set of memory pointers that allow hardware or software to treat the bit vector as a rectangular array. At runtime, the pointer arrays (table lookup addressing) may be used to allow the permutation to be rapidly implemented according to a predetermined pseudo randomization. Likewise, bits along columns can be efficiently accessed using pointer arrays that point to the column elements of each column. That is, the array structure of the constrained interleaver is a mathematical concept and may be implemented in various efficient ways in hardware and/or software. Vectors of pointers can be used to point to rows, to point to elements down a column of an array, or can be used to store a reordering rule for the entire permutation function implemented by an interleaver such as a constrained interleaver. In all cases, table lookup processing is used to speed up interleaver operations for use in real time operation.

The length of the concatenated coded sequence at the output of the IRCC 415 will preferably be M_(SC-IRCC)=(rρn+η)/R_(i), where η is the memory size of the inner code, R_(i) is the rate of the IRCC, and η number of bits are added at the end of each rρn—length frame to terminate the frame, i.e. to force the final state of the IRCC to the zero state. The subscript SC-IRCC refers to a serially concatenated (SC) code that uses IRCC as shown in FIG. 4. It should be noted that the overall rate of the SC-IRCC output from the encoder 415 will be approximately (ignoring the effects of ε) R=R₀*R_(i) where R₀=k/n which is the rate of the (n, k) outer code of the outer encoder 405 and R_(i) is the rate of the IRCC.

The above paragraph describes an SC-IRCC encoding operation that uses constrained interleaving. This encoding can be implemented independently of the mapping operation described below. In a transmitter embodiment, the M_(SC-IRCC) output bits from the inner encoder 415 are additionally sent to a mapper 420 which can be implemented similarly to any of the embodiments of the mapper 320 discussed above. The output of the mapper 420 is an SC-IRCC transmitted signal. FIG. 7 describes transmitters, receivers, and systems that make use of either the SC-BC or SC-IRCC transmitted signals as generated by respective the mapper 320 or 420.

As previously mentioned, the inner code can be selected to be a trellis code which corresponds to a non-recursive convolutional code (possibly a multidimensional trellis code) and the mapper 320 can be selected, for example to be a QAM mapper. In such cases, the outer code 305 and the constrained interleaver 310 can be selected to produce an improved form of trellis coded modulation. While a trellis coded modulation may be improved by using the target trellis code as the inner code in the inner encoder 315, and designing the transmitter 300 to improve the performance of this trellis code, it may be more desirable to instead build an improved trellis coded modulation scheme with a different inner code. For example, it is known that SC-IRCC performs better than serial concatenation with non-recursive convolutional codes. Therefore, the transmitter 400 can be used with a QAM mapper for example to generate a new coded modulation scheme that uses a selected SC-IRCC instead of a non-recursive trellis code. This modified coded modulation signal can be sent over a channel or mapped onto one or more subcarriers in an OFDM embodiment. The design and implementation of such coded modulation schemes using the transmitter apparatus 400 is contemplated for certain embodiments of the present invention.

The general structure of serial concatenation discussed herein, as illustrated in FIGS. 2-4, the serial concatenated encoder includes an outer encoder followed by an interleaver, followed by an inner encoder, which is then followed by a modulator. The modulator is commonly assumed to be memoryless in that the action of the modulator is often to map a codeword onto a constellation point. However, it should be noted that there also exist in the art state-based modulators. State based modulators can be decomposed into an encoder followed by a memoryless modulator. For example, it is well known to those of skill in the art that a continuous phase frequency shift keyed (CPFSK) modulator can be modeled as a continuous phase encoder (CPE) followed by a memoryless modulator. This same concept applies to other more general forms of continues phase modulation (CPM). That is, all CPM modulators can be decomposed into a CPE followed by a memoryless modulator. Further, an encoded form of CPM, that includes an encoder followed by a CPM modulator can also be represented by a composite encoder, which is the equivalent code corresponding to the combination of the code and the CPE, followed by a memoryless encoder.

With that observation, the general structure in FIG. 2-4, which included an outer encoder, followed by an interleaver, followed by an inner encoder, that is then followed by a memoryless modulator, can also be applied to schemes that employ state-based modulation techniques (i.e., modulators with memory). Specifically, the discussion of serial concatenations can also be applied to the following cases; (a) an outer code followed by an interleaver followed by a state-based modulator, and (b) an outer code followed by an interleaver, inner code, followed by a state-based a modulator. In both of these cases, the state-based modulator can be modeled, for example, as a CPE encoder connected in serial with (followed by) a memoryless modulator. In general, other types of state-based modulators other than CPM modulators could be used, so that other types of encoding rules other than CPEs could be used in the state-based modulator. However, a number of preferred embodiments can be envisioned using various CPEs and such designs can advantageously provide constant envelope properties which are often desired in many types of communication systems.

For example, one form of CPM is Minimum Shift Keying (MSK). MSK can be generated using a CPE followed by a memoryless modulator. In the case of MSK, the CPE is given by the rate-1 accumulator encoding rule: y_(k)=[y_(k-1)+x_(k)] mod 2, and a memoryless modulator which is a modulator that selects one of four possible state-transition waveforms as a memoryless function, s(t)=F(y_(k), x_(k), t), over any k th interval, (k−1)T<t≦kT, where T is the signal interval and F(∘, ∘, t) is chosen to supply one of the four known MSK state transition waveforms. Hence, a serial concatenation of an outer code followed by an interleaver, followed by the rate-1 accumulator encoder as described above, which is then followed by the memoryless modulator described above will have the same performance as a serial concatenation of the same outer code followed by the same interleaver, and then followed by a conventional state-based MSK modulator. It should therefore be understood that the present invention applies to embodiments where the inner encoder of serial concatenated encoding systems is inherently implemented as a part of a state-based modulator. For example, blocks 315 and 320 and blocks 415 and 420 can be merged and implemented as a state-based modulator such as an MSK modulator or more generally a CPM modulator, or other types of state-based modulators. The advantage of using constrained interleaving as discussed herein with the general serial concatenation structures are valid with configurations that employ equivalent modulation schemes with memory. Hence, in applications where it is desired to have faster spectral roll-off and a constant envelope, the present invention can be implemented with CPM type modulators as described above.

Moreover, in the case of MSK, because the minimum Euclidian distance of MSK signals and the minimum Euclidian distance of BPSK signals are the same, if the memoryless modulator described above is replaced by a BPSK modulator, the implementation will be simpler and the performance will be identical. Different but similar equivalencies also exist with other forms of CPM and memoryless modulators. Hence all such variations should be understood to be design choices when designing a particular embodiment of the present invention.

FIG. 5 shows a receiver method and apparatus for a receiver 500 used to receive and decode a signal r(t) which was generated in accordance with either of FIG. 3 or FIG. 4 or any of their variants or equivalents. For example, assuming one of the mappers 320 or 420 were used and the transmitted signal was transmitted across a communication channel, then the signal r(t) represents the received version of the transmitted signal as observed at the receiver 500. Block 505 processes or otherwise demodulates r(t) to generate an initial vector r_(S), which preferably corresponds to a vector of bit metrics. As is known in the art, a bit metric is a logarithm of a ratio of the probability that a given bit is a one divided by the probability the same bit is a zero. The length of the vector r_(S) is M_(SC-BC) when r(t) is originated from the transmitter or encoder of FIG. 3, and is of length M_(SC-IRCC) when r(t) is originated from the transmitter or encoder of FIG. 4. Note that if non-binary modulation such as QAM is being used, each symbol will de-map to a given set of bits, each of which will be represented by their respective bit metrics in the vector r_(S). The bit metrics are preferably used by the component codes for a-posteriori probability (APP) decoding.

The portion of the receiver 500 minus the demodulator block 505 corresponds to a decoder structure which may be implemented or used independently of the demodulator block 505. In pure decoder method or apparatus embodiments which are contemplated herein, the receiver 500 minus the block 505 is referred to as the decoder 500. Any discussion herein of the receiver 500 that does not explicitly involve the block 505 also describes the decoder 500 for embodiments where just a decoder is implemented.

The receiver 500 is preferably configured as follows. The receiver block 505 can include any combination of a demodulator, signal conditioning, and bit detector of any variety, to include a soft bit detector that provides bit metrics as are known in the art. However, an aspect of the present invention is to implement the receiver block 505 using shorter block lengths than can be used by prior art systems. For example, if a current standard requires a length N=2000 interleaver, the equalizer, channel estimator, or joint channel estimator/decoder would need to operate on data blocks on the order of length N=2000. If an SC-BC or SC-IRCC is used with a constrained interleaver can achieve the same bit error rate performance, then the equalizer, channel estimator, or joint channel estimator/decoder used in block 505 would be able to operate on data blocks, for example, on the order of length N=200. Likewise, any decision-directed loops in the receiver block 505 would be implemented with the shorter block length. Decision directed loops include a decision feedback equalizer, or decision directed timing recovery loops, for example.

The output of the block 505 couples to an inner code soft in soft out (SISO) decoder 515 for soft decoding and a constrained deinterleaver 510. The inner code soft decoder 515 implements a known soft decoding algorithm such as the BCJR algorithm, a soft output Viterbi algorithm (SOVA), or uses a soft decoding method available for the decoding of LDPC codes, for example. Such algorithms are known to generate extrinsic information which is indicative of the reliability of the soft decoded results. If the soft decoder 515 (or 525) involves an iterative soft decoder like the BCJR algorithm, then one forward and one backward pass through the BCJR algorithm is made for each pass through the overall iterative decoder structure 500. If the soft decoder 515 (or 525) is an LDPC decoder, then as discussed below, it may be desirable to only run one LDPC iteration between variable and check nodes instead of multiple LDPC iterations per pass through the overall iterative decoder 500.

The inner code soft decoder 515 couples its extrinsic information output to a constrained deinterleaver which deinterleaves the extrinsic information received from the inner code soft decoder 515. An outer code soft decoder 525 is coupled to receive the deinterleaved extrinsic information from the constrained deinterleaver 520 and the deinterleaved bit metric (or other type of sample) sequence from the constrained deinterleaver 510. The outer code soft decoder 525 also implements a known soft decoding algorithm such as the BJCR algorithm, the SOVA, or an LDPC decoder, for example. In general, the same or different soft decoding algorithms can be used in the blocks 515 and 525; however the block 515 will operate to soft decode the inner code while the block 525 will operate to soft decode the outer code. The outer code soft decoder 525 couples its output extrinsic information to a stopping criterion block 530. If the stopping criterion block 530 determines that another iteration is needed, the outer code soft decoder 525 also couples its output extrinsic information to a constrained interleaver 535. The output of the constrained interleaver 535 is coupled as an input to the inner code soft decoder 515. If the stopping criterion block 530 determines that another iteration is not needed, then the outer code soft decoder 525 outputs the decoded output sequence and iterations are halted.

The receiver 500 and the decoder 500 operate slightly differently depending on whether the coding is performed as SC-BC or SC-IRCC, i.e., according to FIG. 3 or FIG. 4 respectively. In both cases, the implementation of the soft decoders 515 and 525 and the stopping criterion checker 530 can be implemented using prior art methods so the detailed operation of these blocks is not described herein as it is well known to those skilled in the art. Because the operation of the receiver 500 is different for the SC-BC and SC-IRCC cases, the operation of each case is described separately below.

For the case where receiver 500 is configured to decode an SC-BC, i.e., the inner code is a block code, the soft decoding of the inner code in the block 515 may be performed according to the following actions: 1) Arrange the received symbols in a (n×m) array by feeding the received samples or metrics along columns. Soft decode each column separately using the inner code. All m columns can optionally be decoded in parallel to speed up the decoding. For each of the q message bits of each n-codeword, the soft decoding process will generate q elements of extrinsic information. Once the soft decoding of all m codewords is complete, a (q×m) array of extrinsic information that corresponds to the constrained interleaved bits will available. It should be noted that different memory organizations such as list structures and other data structures may be used that hold the interleaved locations of m n-bit inner codewords. All such equivalent data structures are contemplated, but to keep the discussions herein focused, the array implementation is described herein by way of example.

For the case where receiver 500 is configured to decode an SC-IRCC, i.e., the inner code is an IRCC, the inner code can be soft decoded by feeding in all of the M_(SC-IRCC) bit metrics into the inner decoder 515. Soft decoding of the IRCC can be done by either using the BCJR of the SOVA or some other soft decoding algorithm as discussed above. After the decoding of the inner code, an r×ρn array of extrinsic information corresponding to the constrained interleaved bits will be available. The decoded extrinsic information may be mapped explicitly (or implicitly via memory indirection) to the r×ρn array in column-major order. Alternatively, an output vector of extrinsic information may be left in vector form and inverse-permuted as described below.

The constrained deinterleavers 510, 520 perform the inverse operation of the respective constrained interleaver 310 or 410 or 535 depending on whether an SC-BC or an SC-IRCC is being decoded in the receiver 500. Once the constrained interleaver is constructed as discussed in connection with FIG. 8 and FIG. 13, the constrained interleaver can be viewed as having applied a particular permutation function that is constrained to preserve certain coding properties as discussed below. That is, the constrained interleavers 310 and 410 merely rearrange the bits of the length M_(SC-BC) or M_(SC-IRCC) vector of bits. The constrained deinterleaver then simply performs the inverse permutation that was performed by the corresponding constrained interleaver. If a bit vector X is permuted to a bit vector Y by the constrained interleaver, then any of the deinterleavers 510, 520 rearrange the bits in vector Y to restore the original bit ordering of the vector X. As discussed above, once the permutation is known, such permutation functions and inverse permutation functions can be implemented using lookup tables that can be incremented through to access a sequence of pointers that directly provide the desired reordering rule.

Next the outputs of the deinterleavers 520 and 510 are coupled to the outer code soft decoder 525. The outer code soft decoder 525 uses the de-interleaved extrinsic information and the de-interleaved received bit metrics to soft decode the outer code. Again, a standard known soft decoding algorithm like the BJCR algorithm or the SOVA can be used. When decoding a SC-BC, the codewords of the outer code can be soft decoded by individually decoding each of the m outer block codewords separately. Optionally these individual block codewords can be decoded in parallel to speed up the decoding. If the outer code is an IRCC, the outer code can be soft decoded by processing the received metrics or samples and the soft information coupled to the outer code soft decoder 525 by the deinterleavers 510 and 520. The soft decoding performed by the outer code soft decoder can be based upon the BJCR algorithm, the SOVA, or some other known soft decoding algorithm. In both the SC-BC and SC-IRRC cases, the extrinsic information output of outer code soft decoder 525 is analyzed in block 530 to see if a convergence/stopping criterion has been met. If the stopping criterion has been met, decoding stops and the decoded output sequence is coupled to an output from the outer code soft decoder 525.

If the stopping criterion has not been met, the extrinsic information output of the outer code soft decoder 525 is then constrained interleaved again in block 535 using the respective type of constrained interleaving as described in connection with FIG. 8 and FIG. 13, depending on whether the receiver 500 is decoding an SC-BC or an SC-IRCC. The above decoding process is repeated until convergence is met or until it reaches the highest allowable number of iterations. It is noted that the deinterleaving operation 510 need only be performed on the first iteration of this decoding process because the sequence r_(S) does not change from one iteration to the next.

FIG. 6 illustrates an alternative decoding method 600 and a decoding apparatus 600 for efficiently decoding a constrained-interleaved SC-IRCC where the outer code is a block code similar to the embodiment shown in FIG. 3. The block 505 can be added to the decoder 600 to create a receiver 600. The embodiment of FIG. 6 is based on the list Viterbi algorithm (LVD) which is less computationally complex to implement than the BJCR algorithm or many other iterative decoding algorithms used for serially concatenated codes. List Viterbi decoding is well known to those skilled in the art, see for example: [16] N. Seshadri and C.-E. W. Sundberg, “List Viterbi decoding algorithms with applications,” IEEE Trans. on Commun., vol. 42, pp. 313-323, February/March/April 1994. Also see the references cited therein. Therefore, the detailed implementation of the LVD itself will not be described herein.

Returning to FIG. 6, an initial metrics sequence estimate is output from a receiver portion like the block 505 of FIG. 5. The initial metrics may be measured at the bit or symbol level and measure distance away from a set of constellation point values which can be binary or M-ary in general. The initial metrics are fed to an LVD block 605. The LVD 605 is preferably configured to sequentially output an ordered list of probable decoded sequences starting from the most probable sequence which is the decoded sequence that would be obtained from normal Viterbi decoding. The LVD output sequence is then sent through a constrained deinterleaver 610 and then coupled to an outer block code based match detector 615 where it is used to check to see if the current sequence generated by the LVD corresponds to valid codewords of the outer code. If the current sequence does not match the outer code, then the next sequence in the list is produced by the LVD and is similarly checked until a match is found or a maximum list length is reached. The checking of some or all rρ codewords can be optionally done in parallel. If no match is found, the sequence with the lowest error metric is selected to be the decoded output sequence. An alternative embodiment is to allow the LVD 605 to output a list of sequences and to then use the block 615 to identify the most probable list sequence using the outer code matching process described above.

Again consider trellis coded modulation embodiments where the transmitter 320 is implemented with an inner code that corresponds to a non-recursive trellis code. Many prior art trellis coded modulation systems use a non-recursive convolutional code (in many cases a multi-dimensional trellis code. It is noted that a traditional trellis code is a nonrecursive convolutional code followed by a mapper similar to the mapper 320. These codes can be analyzed in terms of their Euclidean distance. In presence of a mapper a serially concatenated code produced by the transmitter 300 where the inner code corresponds to a non-recursive trellis code can be analyzed in terms of Euclidean distance instead of the Hamming distance. Such an analysis is presented towards the end of this patent application. The transmitter 300 can be implemented with a non-recursive trellis code in the inner encoder 315 with the mapper 320 to generate a modified type of trellis encoded modulation in accordance with an aspect of the present invention. This can significantly improve the bit error rate performance of known trellis codes (including multi-dimensional codes), because the outer code 305 and the constrained interleaver 310 can be configured to improve the minimum Euclidian distance of the concatenation while keeping the error coefficient low as is discussed in further detail below.

For example, if a single parity check (SPC) code is used in the outer code with a known trellis code in the inner encoder 315, the minimum Euclidean distance can be increased by a factor 2 thereby targeting a performance gain close to 3 dB (a more powerful outer code could be used if further gain is required or desired, or to enable a simpler trellis code to be selected as the inner code). The actual gain can be lower due to the reduction in the rate and increase in the error coefficient, but this loss will be made small by the use of the constrained interleaver of a relatively. small length. In order to achieve this performance improvement the receiver can be configured in accordance with either FIG. 5 or FIG. 6 or some variation or equivalent thereof. If the receiver is implemented as per FIG. 5, the receiver will soft decode the trellis code and the outer code and run iterations between the two decoders. If the receiver is implemented as per FIG. 6, the receiver will use LVD decoding of the inner trellis code and find the list item that matches the outer code.

In connection with FIG. 5, a first decoder method and apparatus has been presented that uses iterative decoding using the BCJR algorithm that exchanges extrinsic information between component codes. In connection with FIG. 6, a second decoder approach that uses list Viterbi decoding (LVD) at the inner code and selects the most likely list item that satisfies the parity check equations of the outer code has also been presented.

Additionally, a hybrid of the above two decoders can be constructed in accordance with an aspect of the present invention to provide a third family of decoder embodiments. For example, the BCJR iterations can be employed first to allow the component codes to communicate, interact and to begin to converge. After that interaction, based on the extrinsic information available to the inner code at that point, the LVD can be used by computing the branch metrics in LVD using the apriori probabilities provided by the extrinsic information to the inner code. For example, during any k th interval, if the extrinsic information provided by the outer code is L_(e)(k), then the apriori probabilities of the input bit a_(k)ε{1,0}, P[a_(k)=1] and P[a_(k)=0], at the inner code can be written as

${{P\left\lbrack {a_{k} = 1} \right\rbrack} = \frac{{\mathbb{e}}^{L_{e}{(k)}}}{1 + {\mathbb{e}}^{L_{e}{(k)}}}},{{P\left\lbrack {a_{k} = 0} \right\rbrack} = \frac{1}{1 + {\mathbb{e}}^{L_{e}{(k)}}}}$

In this example, the branch metric of the inner code corresponding to any state transition from state i to state j that results from an input bit a_(i,j) and generates a corresponding output coded bit b_(i,j)ε{0,1} can be calculated as

$\begin{matrix} {{M_{i,j}(k)} = {{\ln\;{P\left\lbrack {a_{k} = a_{i,j}} \right\rbrack}} - \frac{\left( {y_{k} - c_{i,j}} \right)^{2}}{2\sigma^{2}}}} \\ {= {{{L_{e}(k)}a_{i,j}} - \frac{\left( {y_{k} - c_{i,j}} \right)^{2}}{2\sigma^{2}} - {\ln\left\lbrack {1 + {\mathbb{e}}^{L_{e}{(k)}}} \right\rbrack}}} \end{matrix}$

where, c_(i,j)=(2b_(i,j)−1) is the coded bit b_(i,j) in bipolar form, y_(k) is the received sample during the kth interval, and σ²=N₀/2 is the noise variance. Since the ln [1+e^(L) ^(e) ^((k))] term of the above equation is independent of the transition, it can be dropped in the metric calculation. The hybrid approach can be preferably implemented with a stopping criterion on the BCJR iterations. This stopping criterion can be any stopping criterion known to those of skill in the art from the decoding literature.

In addition, the present invention contemplates a stopping criterion based on a distance metric which is calculated, at the end of every iteration, as the Euclidean distance between the received signal and the regenerated signal corresponding to the to currently decoded sequence. The iterations can be stopped when the distance metric reaches a steady value. This is similar in ways to a sign change stopping criterion which stops iterations when there are no more sign changes of the log likelihood ratio (LLR) values. However, a novel stopping criterion is to monitor the distance in place of the sign changes. This provides an advantage over the prior art. It is known that iterative decoding converges to a “fixed point.”. If a fixed point is not reached by the time the prior art stopping criterion has been reached, the decoding solution provided by the prior art is likely to have errors and it is advisable to retransmit that frame.

However, when the iterations are about to converge there is generally a sharp drop in the distance metric. Hence, when using the distance metric in a stopping criterion, the algorithm can check to see if the distance metric has made a sharp drop in recent iterations. If such a drop is detected, this is a sign that the iterations can be carried out beyond the nominal maximum allowable number of iterations, because it is likely that a few more iterations will lead to convergence. This option is not available in the sign change criterion, so the stopping criterion based on the distance metric as disclosed herein can lead to improved performance. However, if the distance metric starts to increase when iterations are continued beyond the maximum allowed number of iterations, the frame is discarded and re-transmission is requested.

In this hybrid decoding method and in related computerized apparatus that implements this method, once the stopping criterion is satisfied, the LVD is preferably initiated using the extrinsic information available at the time the stopping criterion is reached. Next, the most likely list item that satisfies the parity check equations of the outer code can be determined and selected. However, if the stopping criterion is not met when the maximum allowable number of iterations is reached, the LVD algorithm can be initiated disregarding the extrinsic information available at that time as in the decoder shown in FIG. 6.

The hybrid method can also be used in a FEC/ARQ format that by requesting retransmissions for frames that do not satisfy the stopping criterion, or if the LVD requires a long list or both. In such applications, the hybrid BCJR/LVD approach can provide a lower retransmission rate and a lower average list length than using LVD alone as in FIG. 6. Also, in place of the hybrid BCJR/LVD types of embodiments described above, it is also possible to use a hybrid BCJR/LS-MAP decoding technique as is known in the art. However, it is likely that the hybrid BCJR/LVD is more suitable when the inner code is a recursive code especially with constrained interleaving. This is because the LVD algorithm, unlike the LS-MAP decoding algorithm, guarantees that the decoded sequence is indeed a valid sequence according to the state structure of the inner code. This is important, especially in case of recursive codes as they can introduce multiple errors at the output for any single error at the input. Further, in order to realize the improvement of constrained interleaving over uniform interleaving it is necessary to ensure that the selected solution is a valid sequence according to the state structure of the inner code. The proposed hybrid BCJR/LVD and the hybrid BCJR/LS-MAP decoding techniques are general techniques that can be applied to any receiver that employs iterative decoding by exchanging extrinsic information among multiple component codes, including parallel concatenation, serial concatenation, and multi-level coding.

FIG. 7 shows a higher level systems architecture 700 into which any of the SC-CI (serial concatenation with constrained interleaving) techniques described herein may be used. A headend system 705 transmits via a downlink channel to user device 710. The user device 710 transmits back to the headend system 705 via an uplink channel. The headend system comprises a protocol stack 720 which includes a physical layer 724. The physical layer or a coding layer just above the physical layer implement SC-BC or SC-IRCC using constrained interleaving in accordance with the present invention. The headend system also may include a control and routing module to connect to external networks, databases, and the like. The headend system also contains a computer control module 729 which comprises processing power coupled to memory. The computer control module 729 preferably implements any maintenance functions, service provisioning and resource allocation, auto-configuration, software patch downloading and protocol version software downloads, billing, local databases, web page interfaces, upper layer protocol support, subscriber records, and the like.

The user terminal 710 similarly includes a physical layer interface 732, a protocol stack 734 and an application layer module 736 which may include user interface devices as well as application software. The user terminal 710 also may optionally include a packet processor 738 which can be connected to a local area network, for example. The user 710 terminal may also act as an IP switching node or router in addition to user functions in some embodiments.

Another type of embodiment replaces the headend system 705 with another user device 710 in which case direct peer-to-peer communications is enabled. In many applications, though, the headend can act as an intermediary between two user devices to enable indirect peer-to-peer communications using the same headend-to/from-user device uplink/downlink architecture illustrated in FIG. 7. Also, a plurality of networked headends may be employed to the same effect, for example, in a cellular communication system (where the headends are implemented as cellular base stations).

In a preferred embodiment as directly illustrated by FIG. 7, at least one of the uplink and the downlink channels is implemented using one or more of the SC-CI family of encoding/modulation/demodulation and decoding schemes. For example, one or more transmitter and receiver structures such as described with FIGS. 3-6 may be used to implement one or both of the physical layer interfaces 724, 732. In some types of embodiments, the PHYS 724, 732 may also include echo cancellation, cross-talk cancellation, equalization, and other forms of signal conditioning or receiver pre-processing. Also, the transmitted data sequences can be chipped sequences that result by point-wise multiplying bipolar data sequences by bipolar spread spectrum pseudorandom noise type sequences. For example, this could correspond to the CDMA mode in the DOCSIS 2.0 specification. Also, the physical layer channel could be a CDMA wireless channel as well. Many current wireless CDMA systems such as 3G cellular systems use Turbo codes like generated using the structure of FIG. 1 or a variant or equivalent. These systems could be improved using the system architecture of FIG. 7 with the SC-CI coding/decoding implemented in the physical layer. Likewise, SC-CI in accordance with the present invention could be implemented on each subcarrier in an OFDM or OFDMA system to improve a technology such as WiMAX. Alternatively, the headend 705 and the user station 710 can be implemented as nodes in a network where the physical layer devices 724, 732 implement a backbone communication connection between nodes. In such embodiments, the backbone communication connection could involve an SC-CI encoded signal transmitted over cable, microwave, optical, or other means.

Another aspect of the present invention contemplated by FIG. 7 is the provision of services by a communication services provider. The communication service provider provides a communication service such as, for example, a cellular communications service to a set of subscribers, a wireless data service, or supplies a backbone optical communication service to support a network such as the Internet. The service provider implements FIG. 7 or any of its variants or equivalents described above. The service provider employs the PHYS 724, 732 in support of the service. In some cases the service also provides the user devices 710 to the subscribers. This allows the service to be implemented more efficiently and economically that was available with prior art coding technologies.

At this point the basic implementation of technology involving constrained interleaving has been described. Still to be discussed is how and why the constraints are determined and what advantages they provide over prior art uniform interleaving. This discussion is technical in nature and is provided below. Several examples are provided along with performance results in order to help understand the underlying concepts.

To understand the benefits and reasoning behind constrained interleaving, again consider FIG. 3 but consider the constrained interleaver 310 to be a standard unconstrained uniform interleaver. As is shown, two general component codes are used to include: a systematic (q, k) outer code with minimum distance d₀, and a systematic (n, q) inner code with minimum distance d_(i). In serial concatenation with uniform interleaving, m codewords of the outer code, or equivalently mq outer coded bits, are uniformly interleaved and fed into the inner code to form a (mn, mk) serially concatenated code. Following the analysis and the notations of [6], the input-output weight enumerating function (IOWEF) of the serially concatenated code can be written as

$\begin{matrix} {{{C\left( {W,H} \right)} = {{\sum\limits_{l = 0}^{N}\frac{{\left\lbrack {A\left( {W,l} \right)} \right\rbrack^{m}\left\lbrack {B\left( {l,H} \right)} \right\rbrack}^{m}}{\begin{pmatrix} N \\ l \end{pmatrix}}} = {\sum\limits_{w}{\sum\limits_{h}{c_{w,h}W^{w}H^{h}}}}}}{{where},}} & (1) \\ {{{A\left( {W,L} \right)} = {1 + {\sum\limits_{i = d_{0}}^{q}{\sum\limits_{u = 1}^{k}{a_{i,u}W^{u}L^{i}}}}}}{and}} & (2) \\ {{B\left( {L,H} \right)} = {1 + {\sum\limits_{j = d_{i}}^{n}{\sum\limits_{v = 1}^{q}{b_{j,v}L^{v}H^{j}}}}}} & (3) \end{matrix}$ are the weight enumerating functions of the outer and inner codes respectively expressed using the sets of coefficients a_(i,u) and b_(j,v) which are inherent to the two respective codes, and N=qm is the size of the interleaver. Assuming BPSK transmission of coded bits over an additive white Gaussian noise (AWGN) channel with power spectral density N₀/2, the bit error probability P_(be) that follows from the IOWEF in (1) is

$\begin{matrix} {P_{be} \leq {\sum\limits_{w}{\sum\limits_{h}{\frac{w}{mk}c_{w,h}{Q\left( \sqrt{\frac{2{RhE}_{b}}{N_{0}}} \right)}}}}} & (4) \end{matrix}$ where E_(b) is the bit energy, R=k/n is the rate of the code, and Q(.) is the standard Q-function.

It follows from equations (1)-(3) that serial concatenation with uniform interleaving has a minimum distance d_(i). That is, the minimum distance of the serially concatenated code is the same as the minimum distance of the inner code alone. Further, the error coefficient in equation (1) that corresponds to the coded weight d_(i) of the concatenation resulting from an interleaver with weight d₀ is

$\begin{matrix} {\frac{\left\lbrack {m{\sum\limits_{u = 1}^{k}{a_{d_{0},u}W^{u}}}} \right\rbrack\left\lbrack {{mb}_{d_{i},d_{0}}H^{d_{i}}} \right\rbrack}{\begin{pmatrix} {mq} \\ d_{0} \end{pmatrix}}.} & (5) \end{matrix}$

Focusing on the dependence on m, it is seen from equation (5) that the error coefficient related contribution from the above term to the error rate in equation (4) is in the form

$\begin{matrix} {\frac{\lambda_{1}}{m^{({d_{0} - 1})}}{Q\left( \sqrt{\frac{2{Rd}_{i}E_{b}}{N_{0}}} \right)}} & (6) \end{matrix}$ which achieves interleaver gain for d₀≧2, where λ₁ is a constant that depends on the inner and outer codes. Therefore, even though the minimum weight of the concatenation is still d_(i), the impact of the minimum weight codewords on the error rate decreases fast with increasing m. This is why traditionally the minimum distance is not the focus when designing serially concatenated codes. Similarly, the term in equation (1) that corresponds to codewords of the concatenation with weight d₀d_(i) that result from codewords of the outer code with weight d₀ is

$\begin{matrix} \frac{\left\lbrack {m{\sum\limits_{u = 1}^{k}{a_{d_{0},u}W^{u}}}} \right\rbrack\left\lbrack {\begin{pmatrix} m \\ d_{0} \end{pmatrix}b_{d_{i},1}^{d_{0}}H^{d_{i}d_{0}}} \right\rbrack}{\begin{pmatrix} {mq} \\ d_{0} \end{pmatrix}} & (7) \end{matrix}$ and its contribution to the bit error probability is

$\begin{matrix} {\lambda_{2}{Q\left( \sqrt{\frac{2{Rd}_{i}d_{0}E_{b}}{N_{0}}} \right)}} & (8) \end{matrix}$ where λ₂ is a constant that is dependent on the inner and outer codes.

It is seen that error coefficient of the contribution from equation (8) on P_(be) cannot be lowered by increasing m or, in other words, the terms with weight d₀d_(i) do not achieve interleaver gain. Hence, the contribution of the codewords with weight d₀d_(i) of the concatenation generated by codewords with weight d₀ of the outer code acts as a lower bound for the overall error probability of the concatenated code with uniform interleaving.

Note further that it is possible to have codewords with weight d₀d_(i) of the concatenation generated by codewords with weight greater than d₀ of the outer code, and, these all achieve interleaver gain. However, the advantage of increasing the interleaver size of the uniform interleaver will ultimately be limited by the codewords with weight d₀d_(i) generated by codewords with weight d₀ of the outer code.

With uniform interleaving, the impact of the term that corresponds to the weight d_(i) of the concatenation in equation (6) decreases the fastest with increasing m, while the impact of the term with the weight d₀d_(i) in equation (8) approaches a lower limit with increasing m. It can be shown that the all terms with weight lower than d₀d_(i) achieve interleaver gain, and further, the impact of m on the multiplicity gradually decreases as the weight increases until the impact disappears when the weight reaches d₀d_(i). As it has been mentioned in the literature [6], there are terms in equation (4) that have higher weights than d₀d_(i) with increasing error coefficients with m which can degrade performance at lower signal to noise ratio (SNR) values and higher values of m.

Summarizing the above observations, it is noticed that with uniform interleaving with shorter interleaver sizes, the performance is dominated by the variation in equation (6) and similar variations with lower weights. However, the impact of these codewords can be lowered by increasing the size of the interleaver. The impact of the term in equation (8) that corresponds to weight d₀d_(i) codewords of the concatenation does not achieve interleaver gain and hence, cannot be lowered by increasing the size of the interleaver. In addition, as discussed in [6], there are other terms with weight higher than d₀ d_(i) that can have increasing error coefficients that can make significant contributions to the error rate at lower to moderate SNR values despite their higher distances.

The above observations suggest that it would be desirable to design interleavers that could eliminate the contributions from equation (6) and similar variations from all other codewords with weight less than d₀d_(i). If this could be done without significantly increasing the error coefficient of equation (8), attractive P_(be) variations which are dominated by equation (8) could result at reasonably low values of m.

Constrained interleaving is designed to achieve good performance at smaller interleaver sizes. Constrained interleaving uses interleaver constraints to ensure that the minimum distance of the concatenated code is maintained at d₀d_(i) which the maximum achievable minimum distance of the concatenation. Disregarding the impact of the error coefficient, this increase in minimum distance would provide a gain of 10 log₁₀ (d₀) dB. The constrained interleaver is further designed to maximize the actual gain in light of the effects of the error coefficients. The minimum distance of the concatenation is maintained at d₀d_(i) by designing the constrained interleaver to ensure that coded bits of every outer codeword are fed into different codewords of the inner code after interleaving. Constrained interleaving removes contributions from all codewords with weight lower than d₀d_(i) in equation (1). In addition, constrained interleaving preferably uniformly randomizes the interleaver among all interleavers that satisfy the above constraint. In this way, constrained interleaving seeks to jointly maximize the minimum distance of the serially concatenated code while at the same time minimizing the error coefficient subject to this constraint. This combined approach allows much shorter interleavers do the job of what traditionally required a much longer uniform interleaver. Such optimizations to not appear to be possible for parallel concatenated codes, thus providing further reason to adopt serially concatenated codes in practical designs where lower coding delays are desired.

Using the constrained interleaver construction as discussed in connection with FIG. 3 and FIG. 8 below, constrained interleaving can be realized by randomly placing the mq coded bits from the outer code in a q by m rectangular array satisfying the constraint that coded bits from any single codeword from the outer code are placed in q different columns. The interleaved array can then be fed to the inner code along columns. Constrained interleaving is easier to implement for values of m that are integer multiples of q. However, with a slight modification, constrained interleaving can also be used with values of m that are non integer multiples of q.

In constrained interleaving, any weight l interleaver generated by a single non-zero codeword of the outer code will have all its l non-zero positions placed in different columns. Hence, the number of interleavers that satisfy this constraint can be found by realizing that it is allowed to select any set of l out of m columns for the l ‘1’s in the interleaver and to place each of these ‘1’s in any position of the selected column. Hence, the number of constrained interleavers with weight l generated by a single non-zero codeword of the outer code that satisfy the constraint can be written as

$\begin{matrix} {N_{l} = {\begin{pmatrix} m \\ l \end{pmatrix}(q)^{l}}} & (9) \end{matrix}$ Compare equation (9) to the corresponding number of weight l uniform interleavers which is

$\begin{pmatrix} {mq} \\ l \end{pmatrix}$ regardless of the number of non-zero codewords of the outer code that generate the weight of the interleaver.

For example, when m=20, q=4 and l=2, the number of constrained interleavers is 3040 while the number of uniform interleavers is 3160. As one can expect the number of constrained interleavers has to be lower than the number of unconstrained uniform interleavers. However, as the above example shows, the ratio of the number of constrained interleavers to the number of uniform interleavers is not much different from unity. This ratio represents the factor by which the error coefficient is degraded. The number of interleavers, which is the denominator of equation (1), determines the error coefficients. The error coefficients of constrained interleaving are only slightly higher than those of uniform interleaving. The degradation in performance by the above ratio is more than offset by the beneficial action of the constraint, that is, by the complete elimination of the problematic lower order terms that dominate the net error coefficient at lower interleaver sizes, m.

The calculation of the number of possible constrained interleavers in the case of multiple non-zero codewords of the outer code is more complicated as it involves consideration of combinations that have multiple ‘1’s in columns. These expressions are presented below for a specific example with a (4,3) single parity check outer code and a (7,4) inner Hamming code (see equations (10)-(26) and the discussion thereof below). It follows from equation (9) and equations (10)-(26) that the difference between the number of possible constrained interleavers and the uniform interleavers is not that significant, and further they both have the same order of dependence on m.

Constrained interleaving can perform significantly better than uniform interleaving at smaller interleaver sizes. A different way to view the benefits is that constrained interleaving with shorter interleaver sizes can approach the best performance uniform interleaving can achieve with very long interleavers. Even though the effects of the size of the interleaver and the delay associated with it are not generally considered in studies in information theory, they are important considerations in practical applications.

Next consider the implementation and operation of the constrained interleaver 310 when used with the (q, k) outer code and the (n, q) inner code to form the (mn, mk) serially concatenated block code in the SC-BC embodiment of FIG. 3. Consider the interleaving of mq coded bits of the outer code for the case when m is an integer multiple of q, that is, where m=ρq. In this case, the coded bits from exactly ρ codewords of the outer code are placed on a given row. The implementation of the constrained interleaver 310 thus guarantees that the maximum achievable minimum distance for the concatenation is preserved. The constrained interleaver is then further designed to maximize the number of possible constrained interleaver combinations given by equations similar to equation (9). In particular, the implementation needs to ensure that bits on every row and any column of the q by m array can be separately uniformly randomized while satisfying the maximum-minimum-distance constraint. Hence, the permutation function implemented by the constrained interleaver can be constrained in accordance with the following three actions:

-   -   1. Feed-in the coded bits of the outer code into the interleaver         array row by row.     -    (Note that each row will have exactly ρ codewords and all coded         bits from any single codeword of the outer code are in different         columns)     -   2. Randomize the contents of each row separately.     -    Rand_Row_(i)=RandRow_(i)(Row_(i)), i=1, 2, . . . q     -    where Rand_Row_(i) denotes the contents of the i th row after         randomizing, and RandRow_(i) denotes the uniform interleaving         operation used to randomize the contents on the i th row.     -   3. Randomize contents of each column separately.     -    Rand_Column_(j)=RandColumn_(j)(Column_(j)), j=1, 2, . . . m=ρq     -    where Rand_Column_(j) denotes the contents of the j th column         after randomizing, and RandColumn_(j) denotes the uniform         interleaving operation used to randomize the contents on the j         th column. The bits are then read out of the constrained         interleaver in column-major order.

The above three actions ensure that coded bits from any single codeword are placed in separate columns, and any coded bit has the freedom to be placed anywhere in the array. Further, the above implementation ensures that rows and columns are completely randomized and thereby provide the maximum possible number of constrained interleavers.

Referring now to FIG. 8, a flow chart for a method 800 is presented that shows the operations performed in order to implement a constrained for a constrained interleaver designed for an SC-BC. The constraint is implemented to force the constrained interleaver's permutation function to rearrange the order of a set of N=qm bits. The method 800 performs the following operations or their equivalents. At 805 a q×m array of bits is arranged and at 810 the array loaded with a set of outer-encoded input bits. The array has q rows and m columns, and the bits are serially loaded into the array in row-major order with ρq-bit outer code words per row, where ρ=m/q. Also at 810, an i^(th) pseudo-random permutation function is applied to each row i, for i=1, 2, . . . q, wherein the i^(th) pseudo-random permutation function pseudo-randomly rearranges the order of the bits in the i^(th) row. At 815 a j^(th) pseudo-random permutation function is applied to each column j, for j=1, 2, . . . m, where the j^(th) pseudo-random permutation function pseudo-randomly rearranges the order of the bits in the j^(th) column. Also at 815, the bits of each pseudo-randomized column is out of the array in column major order.

It is important to understand that the constrained interleaver identifies a set of row and column permutation functions and applies these same row and column permutation functions to each block of data as the method 800 reaches 820 and loops back to 810. That is, the Rand(∘) function is only called the first time through for each row and column to determine the respective row and column permutations, and all subsequent data blocks are processed using this fixed set of row and column permutations determined on the first pass of the algorithm or off line at design time.

It is also important to realize that the method 800 may only be executed at design time and all subsequent passes of input data blocks through the constrained interleaver can use table lookup operations. That is, the overall length-N permutation function implemented by one pass through the method 800 can be hard coded as a stored vector of pointers that are used to implement the permutation function to process actual data blocks in accordance with table lookup processing as described in more detail below.

FIG. 9 illustrates the implementation of the constrained interleaver when q=3 and m=6 by following the above three actions by numbering the positions of the coded bits of the outer code 1 through 18.

It should be noted that any of the constrained interleavers and constrained deinterleavers shown in FIGS. 3-6 can be implemented in various ways. The above implementation of FIG. 8 is presented to mathematically understand how the constrained interleaver conceptually operates in order to implement the constraints that jointly improve performance by maximizing the minimum distance while reducing the adverse effect of the error coefficient. However, the actual implementation of the constrained interleaver in hardware or software at run time would likely be implemented using register-indirect or memory indirect addressing. That is, once the procedure of FIG. 8 has been performed once, the constraints will have been met, and a known permutation function, Rand_Constrained( ), will be known. Let X, YεB^(M) where B^(M) represents a vector space of M—element binary vectors, and Mε{M_(SC-BC), M_(SC-IRCC)}. Thus let Y=Rand_Constrained(X) and X=DeRand_Constrained(Y). Then the permutation functions Rand_Constrained( ) and DeRand_Constrained( ), once known as per FIG. 8 or FIG. 13, can be implemented as simple table lookup operations. Likewise, the accessing of bits along rows or columns of the rectangular array can be done similarly using register or memory indirect addressing, i.e., via table lookup. For example, when decoding codewords stored in columns, the bits will be spread out in the bit vector, and instead of multiplying by the number of rows, it can be more efficient to use prestored addresses to locate the bits along a given column. All such tables (pointer vectors or matrices of pointers) can be implemented as hardware registers in a processor, as pointer vectors in memory, or can be hard coded into digital logic circuits. It is noted that because the constrained interleaver will be much shorter than a uniform interleaver, that such addressing tables become much more efficient to implement. For example, if a 1000 element array was needed in a turbo decoder, a 120-element hardware register array could be implemented with constrained interleaving to achieve roughly the same effect.

In an alternative embodiment, consider the case where m is not an integer multiple of q. In this case, once the m bits of any row are filled there would be remaining coded bits from the last codeword that should be placed in the next row. In order to ensure that coded bits from any single codeword of the outer code are placed in different columns, action 2 listed above needs to be modified just for the remaining bits of the last codeword of the previous row. Specifically, when randomizing the remaining bits of the last codeword of row i in row (i+1), all columns occupied by the coded bits of the last codeword in the in row i should be excluded. This exclusion ensures that all coded bits of the last codeword from the i th row are placed in different columns. This process should be continued when moving from one row to the next throughout the interleaving process. This adjustment is only required for the last codeword of the previous row and not for any other codewords. Due to this added constraint, the expression for the maximum number of constrained interleavers (like equation (9)) derived for values of m which are integer multiples of q are not exactly correct as they were derived assuming that all rows are randomized without any additional constraints. However, as m increases, the impact of the restriction on the last codeword becomes negligible. In order to keep the constrained interleaver design simple and to maintain the highest possible number interleavers, it is often preferable to employ values of m that are integer multiples of q, which in practice is not too difficult to enforce.

Next focus on a particular exemplary embodiment of the SC-BC of FIGS. 3, 5 and 8. In this example, consider the serial concatenation of an outer (4,3) single parity check (SPC) code with an inner (7,4) Hamming code considered in [6] to generate a (7 m, 3 m) rate 3/7 concatenated code. The concatenation of these codes with uniform and constrained interleaving can be compared by following the above analysis with the respective weight enumerating functions of the selected specific outer and inner codes A(W,L)=1+3WL ²+3W ² L ² +W ³ L ⁴  (10) B(L,H)=1+L(3H ³ +H ⁴)+L ²(3H ³+3H ⁴)+L(H ³+3H ⁴)+L ⁴ H ⁷.  (11)

Following equations (1), (10) and (11), the IOWEF of the concatenated code with uniform interleaving along with an interleaver length N=4 m can be written as

$\begin{matrix} {{C\left( {W,H} \right)} = {1 + \frac{{9{m^{2}\left( {W + W^{2}} \right)}}\mspace{20mu}\left( {H^{3} + H^{4}} \right)}{\begin{pmatrix} {4m} \\ 2 \end{pmatrix}} + \frac{3{m\begin{pmatrix} m \\ 2 \end{pmatrix}}\left( {W + W^{2}} \right)\left( {{3H^{3}} + H^{4}} \right)^{2}}{\begin{pmatrix} {4m} \\ 2 \end{pmatrix}} + \frac{\mspace{11mu}{\left\lbrack {{9\begin{pmatrix} m \\ 2 \end{pmatrix}\left( {W + W^{2}} \right)^{2}} + {m\; W^{3}}} \right\rbrack\left\lbrack {{9\begin{pmatrix} m \\ 2 \end{pmatrix}\left( {H^{3} + H^{4}} \right)^{2}} + \mspace{110mu}{{m\left( {m - 1} \right)}\left( {{3H^{3}} + H^{4}} \right)\left( {H^{3} + {3H^{4}}} \right)}} \right\rbrack}}{\begin{pmatrix} {4m} \\ 4 \end{pmatrix}} + {{other}\mspace{14mu}{{terms}.}}}} & (12) \end{matrix}$

Even though it is not necessary to structure the uniform interleaver as a rectangular array, in order to compare uniform and constrained interleaving, without loss of generality, we assume that the same q by m array structure is used with uniform interleaving. In the case of uniform interleaving, no constraint is applied so that coded bits of the outer code can be randomly placed anywhere in the rectangular array making it equivalent to a uniform interleaver with size N=4 m.

To first analyze the performance of the serially concatenated code with uniform interleaving, note that the first term in equation (12) corresponds to the all zero codeword. The second term corresponds to l=2 with two “1”s in the 4 by m array of coded bits of the outer code with both “1”s located in a single column of the interleaver array. Note also that the second term generates a codeword with minimum weight 3 of the concatenation. The third term of equation (12) corresponds to again l=2 but using two columns of the interleaver resulting in a distance of at least 6. The fourth term corresponds l=4 but still using only two columns and generating codewords with weight 6 and higher. Similar terms that generate codewords with weight 6 are not that significant due to a larger denominator and/or larger weight, as can be found by considering l=6 using only with two columns. Therefore, in this discussion only the first four terms of equation (12) are considered as the primary terms. Starting with equation (12) it can be seen that the coefficients c_(w,h) of the dominant terms in equation (1) can be written as

$\begin{matrix} {{c_{1,3} = {c_{1,4} = {c_{2,3} = {c_{2,4} = \frac{9m^{2}}{\begin{pmatrix} {4m} \\ 2 \end{pmatrix}}}}}};} & (13) \\ {{{c_{1,6} = \frac{27{m\begin{pmatrix} m \\ 2 \end{pmatrix}}}{\begin{pmatrix} {4m} \\ 2 \end{pmatrix}}};}{{{c_{2,6} = {\frac{27{m\begin{pmatrix} m \\ 2 \end{pmatrix}}}{\begin{pmatrix} {4m} \\ 2 \end{pmatrix}} + \frac{{81\begin{pmatrix} m \\ 2 \end{pmatrix}^{2}} + {3{m\left( {m - 1} \right)}}}{\begin{pmatrix} {4m} \\ 4 \end{pmatrix}}}};}.}} & (14) \\ {{{c_{3,6} = \frac{\left\lbrack {{18\begin{pmatrix} m \\ 2 \end{pmatrix}} + m} \right\rbrack\left\lbrack {{9\begin{pmatrix} m \\ 2 \end{pmatrix}} + {3{m\left( {m - 1} \right)}}} \right\rbrack}{\begin{pmatrix} {4m} \\ 4 \end{pmatrix}}};}{c_{4,6} = {\frac{9{\begin{pmatrix} m \\ 2 \end{pmatrix}\left\lbrack {{9\begin{pmatrix} m \\ 2 \end{pmatrix}} + {3{m\left( {m - 1} \right)}}} \right\rbrack}}{\begin{pmatrix} {4m} \\ 4 \end{pmatrix}}.}}} & (15) \end{matrix}$

Considering the contributions from the lower weight terms in equation (1), specifically considering contributions corresponding to terms in equations (13), (14) and (15), P_(be) can be written as

$\begin{matrix} {P_{be} \leq {{\frac{3}{2\left( {{4m} - 1} \right)}{Q\left( \sqrt{\frac{6\; R\; E_{b}}{N_{0}}} \right)}} + {\frac{3}{2\left( {{4m} - 1} \right)}{Q\left( \sqrt{\frac{8R\; E_{b}}{N_{0}}} \right)}} + {\frac{27\left( {m - 1} \right)}{4\left( {{4m} - 1} \right)}{Q\left( \sqrt{\frac{12R\; E_{b}}{N_{0}}} \right)}} + {{other}\mspace{14mu}{terms}\mspace{14mu}{with}\mspace{14mu}{higher}\mspace{14mu}{{distances}.}}}} & (16) \end{matrix}$ The first term of equation (16) with the minimum argument of the Q function corresponds to l=2 using a single column of the interleaver array, second term corresponds to l=4 again using a single column of the array, and the third term corresponds to l=4 using two columns of the array each with weight 2. It is seen that the first two terms in equation (16), decrease with increasing m and achieve interleaver gain. However, the third term in equation (16) does not achieve interleaver gain. In fact the third term of equation (16) can be considered as a lower bound for P_(be) as

$\begin{matrix} {P_{be} > {\frac{27\left( {m - 1} \right)}{4\left( {{4m} - 1} \right)}{Q\left( \sqrt{\frac{12R\; E_{b}}{N_{0}}} \right)}}} & (17) \end{matrix}$

To next analyze the performance of the same serially concatenated code but with constrained interleaving, note that by the design of the constraint, that the first two terms of equation (16) are eliminated. The number of constrained interleavers at different interleaver weights l can then be found by considering all possible combinations of obtaining that value of l subject to the constraint. For example, only a single codeword of the outer SPC can generate l=2 interleavers, and hence, the number of constrained interleavers is

$\begin{matrix} {N_{2} = {{(4)(4)\begin{pmatrix} m \\ 2 \end{pmatrix}} = {8{{m\left( {m - 1} \right)}.}}}} & (18) \end{matrix}$

Using the above analysis, note that with this serially concatenated code, when m=4, uniform interleaving has 120 interleaver combinations and constrained interleaving has 96 combinations. The reduction of (120−96)=24 combinations from uniform interleaving is due to not allowing both “1”s to be in the same column. However, constrained interleaving eliminates the lower order terms that give the highest error coefficient at lower values of interleaver size, m, while at the same time increasing the minimum distance of the serially concatenated code. This provides a net coding gain, especially at shorter interleaver sizes.

The case l=4 when m≧4 can be obtained either from one codeword of weight 4 or from two codewords each with weight 2 of the outer code. When generated from a single codeword of the outer code, all four “1”s will be placed in 4 different columns, and the number of possible constrained interleavers is

$\begin{matrix} {N_{4b} = {\begin{pmatrix} m \\ 4 \end{pmatrix}(4)^{4}}} & (19) \end{matrix}$ and the resulting weight of the concatenation is at least 12. Similarly, case l=4 can also be generated from two different codewords each with weight 2 by placing the four “1”s either in 2 columns (each with two “1”s), or in three columns (one with two “1”s and two with one “1”) or in four columns each (with one “1”), generating a total number of constrained interleavers

$\begin{matrix} {N_{4a} = {{\begin{pmatrix} m \\ 4 \end{pmatrix}(4)^{4}} + {\begin{pmatrix} m \\ 3 \end{pmatrix}(3)\begin{pmatrix} 4 \\ 2 \end{pmatrix}(4)^{2}} + {\begin{pmatrix} m \\ 2 \end{pmatrix}\begin{pmatrix} 4 \\ 2 \end{pmatrix}^{2}}}} & (20) \end{matrix}$ with a minimum weight of at least 6. Hence, it is clear that in constrained interleaving the number of possible interleavers can vary depending on how the weight of the interleaver is generated from the outer code.

Similarly, when l=6 and m≧6, the number of constrained interleavers can be found when it is generated by three weight 2 codewords of the outer code as

$\begin{matrix} {N_{6a} = {{4096\begin{pmatrix} m \\ 6 \end{pmatrix}} + {7680\begin{pmatrix} m \\ 5 \end{pmatrix}} + {4480\begin{pmatrix} m \\ 4 \end{pmatrix}} + {792\begin{pmatrix} m \\ 3 \end{pmatrix}} + {16\begin{pmatrix} m \\ 2 \end{pmatrix}}}} & (21) \end{matrix}$ and when it is generated by one weight 2 and one weight 4 codeword of the outer code as

$\begin{matrix} {N_{6b} = {{4096\begin{pmatrix} m \\ 6 \end{pmatrix}} + {7680\begin{pmatrix} m \\ 5 \end{pmatrix}} + {3456{\begin{pmatrix} m \\ 4 \end{pmatrix}.}}}} & (22) \end{matrix}$

Similarly, when l=8 and m≧8, the number of constrained interleavers can be found when it is generated by four weight 2 codewords of the outer code as

$\begin{matrix} {N_{8a} = {{65536\begin{pmatrix} m \\ 8 \end{pmatrix}} + {172032\begin{pmatrix} m \\ 7 \end{pmatrix}} + {162816\begin{pmatrix} m \\ 6 \end{pmatrix}} + {66560\begin{pmatrix} m \\ 5 \end{pmatrix}} + {10896\begin{pmatrix} m \\ 4 \end{pmatrix}} + {384\begin{pmatrix} m \\ 3 \end{pmatrix}} + \begin{pmatrix} m \\ 2 \end{pmatrix}}} & (23) \end{matrix}$ or when it is generated by one weight 4 and two weight 2 codewords of the outer code as

$\begin{matrix} {N_{8b} = {{65536\begin{pmatrix} m \\ 8 \end{pmatrix}} + {172032\begin{pmatrix} m \\ 7 \end{pmatrix}} + {162816\begin{pmatrix} m \\ 6 \end{pmatrix}} + {65280\begin{pmatrix} m \\ 5 \end{pmatrix}} + {9744\begin{pmatrix} m \\ 4 \end{pmatrix}}}} & (24) \end{matrix}$ and when it is generated by two weight 4 codewords of the outer code as

$\begin{matrix} {N_{8c} = {{65536\begin{pmatrix} m \\ 8 \end{pmatrix}} + {172032\begin{pmatrix} m \\ 7 \end{pmatrix}} + {138240\begin{pmatrix} m \\ 6 \end{pmatrix}} + {34560\begin{pmatrix} m \\ 5 \end{pmatrix}} + {1296{\begin{pmatrix} m \\ 4 \end{pmatrix}.}}}} & (25) \end{matrix}$

The number of interleavers with constrained interleaving can be used in equation (12) and by dropping the terms that are prevented by the constraint, the IOWEF of the concatenated code with constrained interleaving can be expressed as

$\begin{matrix} {{C_{C}\left( {W,H} \right)} = {1 + \frac{3{m\begin{pmatrix} m \\ 2 \end{pmatrix}}\left( {W + W^{2}} \right)\left( {{3H^{3}} + H^{4}} \right)^{2}}{N_{2}} + \frac{\left\lbrack {9\begin{pmatrix} m \\ 2 \end{pmatrix}\left( {W + W^{2}} \right)^{2}} \right\rbrack\begin{bmatrix} {{9\begin{pmatrix} m \\ 2 \end{pmatrix}\left( {H^{3} + H^{4}} \right)^{2}} + {m\left( {m - 1} \right)}} \\ {\left( {{3H^{3}} + H^{4}} \right)\left( {H^{3} + {3H^{4}}} \right)} \end{bmatrix}}{N_{4a}} + \frac{{mW}^{3}\left\lbrack {9\begin{pmatrix} m \\ 2 \end{pmatrix}\begin{matrix} {\left( {H^{3} + H^{4}} \right)^{2} + {m\left( {m - 1} \right)}} \\ {\left( {{3H^{3}} + H^{4}} \right)\left( {H^{3} + {3H^{4}}} \right)} \end{matrix}} \right\rbrack}{N_{4b}} + {{other}\mspace{14mu}{{terms}.}}}} & (26) \end{matrix}$

The new weight enumeration function in equation (26) along with equation (1) determines the error rate bound for serial concatenation with constrained interleaving.

FIG. 10 illustrates how constrained interleaving can reach the performance bound of uniform interleaving, but with a much shorter interleaver. The uniform interleaving curves of FIG. 1 show the variations of the combined effect of the first two terms of equation (16) and the lower bound in equation (17) for different values of interleaver length, N=16, N=40 and N=400, along with the P_(be) upper bound in equation (1) considering all codewords of weight up to 16 along with the error rate variations of constrained interleaving. It is seen from FIG. 10 that unless the uniform interleaver length is very large, the first two terms of equation (16) dominate the performance at moderate to higher SNR values. It is also noticed that P_(be) cannot be lowered below the lower bound by increasing the length of the uniform interleaver, m. On the other hand, FIG. 10 indicates that constrained interleaving can approach the performance bound with a much smaller interleaver. Further, it is seen that at very low error probabilities, below about 10⁻⁷, constrained interleaving begins to perform significantly better than uniform interleaving because of the constraint's ability to overcome the error rate floor effects of uniform interleaving. This improvement in performance would be useful for applications that operate at low bit error rates such as in optical communication systems and in magnetic recording.

As discussed in the background section, SPC codes are well known. For example, it is known that a d-dimensional SPC code with an overall rate of

$\left( \frac{m - 1}{m} \right)^{d}$ and a minimum Hamming distance of 2^(d) can be generated by using (m, m−1) SPC codes along all of the d dimensions. Even though the minimum distance of the code can be increased by increasing the number of dimensions, it also increases the error coefficient of the code. Specifically, the bit error probability of a 2-dimensional (2-D) SPC can be approximately expressed as (see [11] as referenced in the background section herein):

$\begin{matrix} {P_{be} \approx {\frac{\left( {m - 1} \right)^{2}}{2}{Q\left( \sqrt{\frac{8\left( {m - 1} \right)^{2}E_{b}}{m^{2}N_{0}}} \right)}}} & (27) \end{matrix}$

In the literature multi-dimensional SPC codes have been discussed by using uniform interleaving in between dimensions. It is found that interleaving can improve performance of multi-dimensional SPCs when the number of dimensions is above 2, however, in 2-D SPCs uniform interleaving cannot improve performance over the same scheme without interleaving. It is stated in [11] that this is due to the lowering of the minimum distance in case of uniform interleaving. Since both component codes of 2-D SPC have the same weight enumerating function given by

$\begin{matrix} {{D\left( {W,L} \right)} = {1 + {\sum\limits_{{i\;{odd}},{i > 0}}{\begin{pmatrix} {p - 1} \\ i \end{pmatrix}W^{i}L^{i + 1}}} + {\sum\limits_{{i\;{even}},{i > 0}}{\begin{pmatrix} {p - 1} \\ i \end{pmatrix}W^{i}L^{i}}}}} & (28) \end{matrix}$ the weight enumerating function of the serial concatenation of them with uniform interleaving is given by

$\begin{matrix} {{C\left( {W,H} \right)} = {{\sum\limits_{i = 0}^{m{({m - 1})}}\frac{{\left\lbrack {D\left( {W,l} \right)} \right\rbrack^{({m - 1})}\left\lbrack {D\left( {l,H} \right)} \right\rbrack}^{m}}{\begin{pmatrix} {m\left( {m - 1} \right)} \\ l \end{pmatrix}}} = {\sum\limits_{w}{\sum\limits_{h}{c_{w,h}W^{w}{H^{h}.}}}}}} & (29) \end{matrix}$ The lower weight terms (l=2 terms), equation (29) can be written as

$\begin{matrix} {{C\left( {W,H} \right)} = {1 + \frac{{m\left( {m - 1} \right)}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}^{2}W^{2}H^{2}}{\begin{pmatrix} {m\left( {m - 1} \right)} \\ 2 \end{pmatrix}} + \frac{{m\left( {m - 1} \right)}^{2}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}{WH}^{2}}{\begin{pmatrix} {m\left( {m - 1} \right)} \\ 2 \end{pmatrix}} + {\frac{\left( {m - 1} \right)^{3}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}\begin{pmatrix} m \\ 2 \end{pmatrix}}{\begin{pmatrix} {m\left( {m - 1} \right)} \\ 2 \end{pmatrix}}W^{2}H^{4}} + \frac{\left( {m - 1} \right)^{4}\begin{pmatrix} m \\ 2 \end{pmatrix}{WH}^{4}}{\begin{pmatrix} {m\left( {m - 1} \right)} \\ 2 \end{pmatrix}} + {{other}\mspace{14mu}{{terms}.}}}} & (30) \end{matrix}$

It is seen from the second and third terms of equation (30) that uniform interleaving only achieves a minimum distance of 2. Further, it is seen from the second term of equation (30) that, with respect to the error coefficient, the numerator is in the order of m⁶ and denominator is in order of m⁴. Since the size of the interleaver is in the order of m², the second term does not achieve interleaver gain. This explains why 2-D SPC with uniform interleaving cannot perform better than 2-D SPC without interleaving which is given by equation (27).

Next consider 2-D SPC with constrained interleaving. Constrained interleaving of 2-D SPC can be performed by arranging (m−1) codewords of the first dimension, each m bits long, in a (m−1) by m array and interleaving by satisfying the constraint of constrained interleaving as previously discussed. With constrained interleaving, the second and third terms of equation (30) are thereby eliminated. Further, the denominator of the remaining fourth term of equation (30) is modified as

$\begin{matrix} {N_{2} = {\begin{pmatrix} m \\ 2 \end{pmatrix}{\left( {m - 1} \right)^{2}.}}} & (31) \end{matrix}$ Hence, the weight enumerating function of 2-D SPC with constrained interleaving can be derived from equation (29), by also considering the terms that become important after the fourth term in equation (29), as

$\begin{matrix} {{C_{C}\left( {W,H} \right)} = {{1 + {\frac{{m\left( {m - 1} \right)}^{3}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}\begin{pmatrix} m \\ 2 \end{pmatrix}}{\left( {m - 1} \right)^{2}\begin{pmatrix} m \\ 2 \end{pmatrix}}W^{2}H^{2}} + \frac{\left( {m - 1} \right)^{4}\begin{pmatrix} m \\ 2 \end{pmatrix}{WH}^{4}}{\left( {m - 1} \right)^{2}\begin{pmatrix} m \\ 2 \end{pmatrix}}} = {1 + {\left( {m - 1} \right)\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}W^{2}H^{4}} + {\left( {m - 1} \right)^{2}{WH}^{4}} + {{other}\mspace{14mu}{{terms}.}}}}} & (32) \end{matrix}$

Since the interleaver size is in the order of m², the error coefficient of the bit error probability resulting from the second term of equation (32) is in the order of m. However, the rate and hence, the argument of the Q-function also increases with increasing m. Compared with the performance of 2-D SPC without interleaving given by equation (27) that has error coefficient in the order of m², constrained interleaving achieves an interleaver gain on the order of 1/m. The contributions of the “other terms” of equation (32), which should also be considered at low SNR values, can be found by considering terms with N₄, N₆, etc. These values for the (m, m−1) SPC outer code considered in 2-D SPC can be found by modifying equations (19)-(25). For example, using the same notations, equations (19) and (20) would be modified as

$\begin{matrix} {N_{4a} = {{\begin{pmatrix} m \\ 4 \end{pmatrix}\left( {m - 1} \right)^{4}} + {3\begin{pmatrix} m \\ 3 \end{pmatrix}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}\left( {m - 1} \right)^{2}} + {\begin{pmatrix} m \\ 2 \end{pmatrix}\begin{pmatrix} {m - 1} \\ 2 \end{pmatrix}^{2}}}} & (33) \\ {\mspace{79mu}{N_{4b} = {\begin{pmatrix} m \\ 4 \end{pmatrix}{\left( {m - 1} \right)^{4}.}}}} & (34) \end{matrix}$

FIG. 11 shows the error rate bounds of 2-D SPC codes with uniform interleaving and constrained interleaving along with that of without interleaving when the interleaver lengths are N=12 and N=90 by considering terms with weights up to 8 in the interleaver. Since the rate of the code varies with m, in order to observe the impact on interleaver length, the error rate variations when the rate r is fixed at 1 are also plotted. The results of FIG. 11 indicated that 2-D SPC codes can benefit significantly from constrained interleaving compared with both uniform interleaving and without interleaving.

Constrained interleaving can also improve performance in 3-D and higher dimensional SPCs as well. In case of 3-D SPCs, (m−1) separately constrained interleaved 2-D SPC coded bits (each with m(m−1) bits) are arranged in a (m−1) by m (m−1) array and are interleaved again according to constrained interleaving. By extending the analysis of 2-D SPC with constrained interleaving, it is possible to show that even with 3-D SPC (and similarly higher dimensional SPC), constrained interleaving can perform better than uniform interleaving.

At this point the SC-BC implementations and the theory behind them have been described. We now turn our attention to the SC-IRCC case. The analysis of convolutional codes differs from that of block codes due to the absence of a well defined block length. However, convolutional codes can be analyzed using the concept of equivalent block codes. Specifically, a rate R=p/n convolutional code with memory v can be analyzed by considering its equivalent (N/R, N−pv) block code and considering all of the N input bits (including the pv termination bits) as in a single block. The weight enumeration function (WEF) of the concatenated code can be found by considering all possible error events and their concatenations within the block of N input bits.

FIG. 4 shows an embodiment of an SC-IRCC that uses a constrained interleaver 410 and an outer block code 405 which could be alternatively implemented by a non-recursive convolutional. In FIG. 4, the inner code 415 is implemented as an IRCC. In order to understand the advantages of the constrained interleaving with convolutional codes, we first review the analysis of SCCC (serial concatenation with convolutional codes) with uniform interleaving as presented in [6]. The observations with uniform interleaving are then used to motivate and develop constrained interleaving for use with SC-IRCC. That is, in FIG. 4, let us assume that no constraints are applied and the constrained interleaver 410 is an (unconstrained) uniform interleaver.

Adopting the same notation in [6], the performance of a SCCC over an additive white Gaussian noise (AWGN) channel with power spectral density N₀/2 can be bounded by considering all error events of different weights and their contributions to the bit error probability P_(be) using the union bound. The resulting P_(be) of the concatenated code can be bounded in terms of the standard Q-function as

$\begin{matrix} {P_{be} \leq {\sum\limits_{k}{{c\left( h_{k} \right)}N^{\alpha{(h_{k})}}{Q\left( \sqrt{\frac{2{Rh}_{k}E_{b}}{N_{0}}} \right)}}}} & (35) \end{matrix}$ where h_(k) is the weight of the k th error event of the concatenation, α(h_(k)) is the exponent of the interleaver size N, c(h_(k)) is a constant dependent on the component codes and the weight h_(k) but not on N, R is the rate of the code, and E_(b)·/N₀ is the signal to noise ratio. The analysis of the performance in equation (35) focuses on the weights h_(k), that determine the argument of the Q function, and their respective exponents, α(h_(k))), that determine the error coefficient and the interleaver gain of the respective weight h_(k). The minimum value of h_(k), h_(m), of the concatenated code can be higher than the minimum free distance of the inner code, d_(f) ^(i), due to the influence of the outer code. Further, as stated in [6], generally, h_(m)<2d_(f) ^(i). This implies that the distance of the outer code can sometimes improve the distance of the prior art serially concatenated codes in accordance with d_(i)<d_(sc)<2d_(i). However, this has nothing to do with the permutation function chosen for the interleaver, but has to do instead with the selection of the inner and the outer codes.

Serial concatenation of convolutional codes with uniform interleaving has been analyzed by considering: (a) the values of h_(m) and its corresponding value of the exponent α(h_(m)), and (b) the maximum exponent of any weight, α_(M)=Max {α(h_(k))} [6]. The analysis of (a) determines the error rate variation in equation (35) at high SNR values while the analysis of (b) determines the impact of the interleaver size on any weight and checks if α(h_(k)) is negative for all values of h_(k), thereby guaranteeing interleaver gain for all weights. Related to the analysis of (a), it has been shown that [6] α(h _(m))≧1−d _(f) ⁰  (36) and hence, the minimum weight term achieves interleaver gain for d_(f) ⁰≧2, where d_(f) ⁰ is the minimum Hamming distance of the outer code. Related to the analysis of (b), it has been shown that the α(h_(k)) value corresponding to any weight h_(k) is given by [6] α(h _(k))=n ₀ +n _(i) −l−1  (37) where n₀ and n_(i) are the number of error events concatenated on the trellises of the outer and inner codes respectively corresponding to a weight l interleaver that generates the weight h_(k) sequence at the output of the concatenation. It is seen from equation (37) that the maximum α(h_(k)) value results from maximum possible n₀ and n_(i) values at any given value of l. The important observations related to equation (37) are listed below. (i) The interleaver weight l≧n₀d_(f) ⁰. (ii) In case of block or non-recursive inner convolutional codes, maximum n_(i)=l, or n_(i)<l. (iii) Since the input weight of an error event of an IRCC is at least 2, when the inner code is an IRCC, the maximum value of n_(i) is l/2 for even l, and [(l−3)/2+1] for odd values of l. Further, the minimum weight of the output of the IRCC with maximum n_(i) requires the minimum weight of the error event generated by a weight 2 input of the inner code which is referred to as the effective minimum weight d_(f,eff) ^(i), and also by the minimum output weight of the inner code corresponding to a weight 3 input, h_(m)(3).

With the above observations, it follows from equation (37) that inner block or non-recursive convolutional codes can have weights h_(k) with positive α(h_(k)) values and hence, their contributions to the error rate in equation (35) increase with increasing interleaver size [1, 2]. It is also seen that with recursive inner codes, when d_(f) ⁰≧2, α(h_(k)) is always negative guaranteeing interleaver gain for all weights h_(k) in equation (35). Specifically, for IRCCs [6],

$\begin{matrix} {\alpha_{M} = {- {\left\lfloor \frac{d_{f}^{0} + 1}{2} \right\rfloor.}}} & (38) \end{matrix}$ Hence, IRCCs are better than block or non-recursive convolutional codes when used as inner codes in serial concatenation with uniform interleaving [6]. Also, it is desirable to use an outer code with a higher, and preferably an odd, free minimum distance d_(f) ⁰. Further, the weight h_(k) that corresponds to α_(M), denoted by h(α_(M)), is given by [6]:

$\begin{matrix} {{h\left( \alpha_{M} \right)} = \left\{ \begin{matrix} {{d_{f}^{0}d_{f,{eff}}^{i}\text{/}2},} & {d_{f}^{0}{even}} \\ {\left\lbrack {{\frac{\left( {d_{f}^{0} - 3} \right)}{2}d_{f,{eff}}^{i}} + {h_{m}(3)}} \right\rbrack,} & {d_{f}^{0}{odd}} \end{matrix} \right.} & (39) \end{matrix}$

Finally, it is known that the outer code of a SCCC should preferably be a non-recursive convolutional code and not a recursive code, and also it is known that the behavior of block codes and non-recursive codes are similar when used as outer codes in serial concatenation[6].

The above observations about the bit error rate performance of SCCCs that use uniform interleaving of SCCC are next used to develop constrained interleaving techniques that operate with SC-IRCCs. Since block codes and non-recursive convolutional codes behave the same way, and since constrained interleaving is easier to implement with block outer codes, without loss of generality, for now we focus on SC-IRCCs embodiments that use an outer (n,k) block code with an IRCC as illustrated in FIG. 3. Due to the absence of a lower bound as in case of uniform interleaving with serially concatenated block codes, the goal of constrained interleaving of SC-IRCC is to improve performance over uniform interleaving at any given interleaver length, but not to try to approach any lower bound as was done with inner block codes. Further, due to the absence of fixed block lengths of convolutional codes, a different set of interleaver constraints are needed to optimize or improve the performance of SC-IRCCs.

Similar to constrained interleaving of SC-BCs, let us consider constrained interleaving of SC-IRCC in the form of a r by ρn array and feeding interleaved bits, with u termination bits, into the inner code along columns as illustrated in FIG. 12. As shown in FIG. 12, this array holds the same information as a vector of rρn input bits (or bit metrics or extrinsic values). We consider the number of columns of the interleaver array to be an integer multiple of n, where as indicated above, n is the number of bits in each codeword. Hence, rρ number of codewords of the outer code will be fed into the interleaver with ρ codewords placed along each row. As with block codes, the objective here is to maintain the minimum weight of the concatenation at d₀d_(f) ^(i), where d₀ is the minimum distance of the outer block code (since the outer code is a block code in this discussion, we simply denote the minimum distance of the outer code by d₀ instead of) d_(f) ⁰). In order to achieve this objective, it is necessary to ensure that each coded bit of the outer code gains at least a weight of d_(f) ^(i) when the interleaved bits are passed through the inner code. Hence, due to the trellis structure of the inner convolutional code, it is necessary to maintain enough spacing between the coded bits of each codeword of the outer code to maintain the minimum distance of the concatenation when the outer code is fed into the inner code. This suggests that the easiest way to constrained interleave SC-IRCC is to place all coded bits of the any codeword of the outer code along the same row of the interleaver.

Even though this prevents randomizing the contents of columns that was allowed with block inner codes, randomizing within rows and shuffling of the rows can still be allowed with IRCC. The value of r, that depends on the inner code, should be selected to maintain the minimum distance of the concatenation at d₀d_(f) ^(i). The constrained interleaver of SC-IRCC can be implemented by placing rρ number of codewords of the outer code into an Input_Block, applying uniform interleaving at the codeword level to the n-bit codewords in the Input_Block, and placing the randomized codewords into a length rρ vector of codewords, Rand_Input_Block. The memory structure is organized to then consider the Rand_Input_Block to be an r×nρ rectangular array of bits which constitutes the constrained interleaver array. A vector of r-element row pointers, *Rows, can be constructed where the i^(th) element of *Rows, points to the beginning of the i^(th) row of the constrained interleaver array. This allows the Rand_Input_Block to be manipulated in hardware or software as an r×ρn rectangular array of bits.

The constrained interleaver can be implemented or its permutation function can be designed by taking the actions summarized below:

-   -   1. Randomize the length-rρ Input_Block of codewords (CW's).     -   Rand_Input_Block=Rand_CW (Input_Block),     -   where Rand_Input_Block denotes a uniformly interleaved set of         n-bit codewords of the outer code after randomizing, and Rand_CW         denotes the uniform interleaving operation applied to randomize         n-bit codewords as opposed to bits.     -   2. Randomize the contents of each row separately.     -   Rand_Row_(i)=RandRow_(i)(Row_(i)), i=1, 2, . . . r     -   where Rand_Row_(i) denotes the contents of the i th row after         randomizing, and RandRow_(i) denotes the uniform interleaving         operation used to randomize the contents of the i th row.

The bits are then read out of the interleaver in column-major order. It is noticed that by following the above actions, any codeword of the outer code has the freedom to be placed in any row, codewords have the freedom to get mixed up randomly, and coded bits of any codeword get placed along the same row of the interleaver thereby ensuring the highest possible minimum distance of d₀d_(f) ^(i) while maximizing the number of interleavers.

FIG. 13 illustrates the operations of a constrained interleaver 1300 designed to implement a constraint to jointly take into consideration the minimum distance and error coefficients of an SC-IRCCs. The constraint is implemented to force the permutation function rearrange the order of a set of N=rρn bits to be equivalent to performing a set of operations as described below. At 1305 a set of parameters as discussed below are determined for implementation of the constrained interleaver. A rectangular array data structure is configured, preferably using a vector of row pointers to implement row addressing and row swapping more efficiently. At 1310 an input block of outer encoded bits is formed. Also at 1310 a codeword-level permutation function is applied to randomize an ordering of rρ number of n-bit outer code words embedded in the input block. The rρn number of outer-encoded bits from the input block are loaded into a r×ρn array of bits, wherein the array has r rows and ρn columns, and the bits are serially loaded into the array in row-major order with ρ number of n-bit outer code words per row. Next at 1315 an i^(th) pseudo-random row permutation function is applied to each row i, for i=1, 2, . . . r, wherein the i^(th) pseudo-random permutation function pseudo-randomly rearranges the order of ρn coded bits in the i^(th) row. At 1320 the bits are read out of the array in column major order.

Similar to the discussion made in connection with FIG. 8, if the same pseudo randomized permutation functions are used in each pass, then at step 1325 a new block of data is brought in and the constrained interleaving is repeated on the next input data block using the same set of codeword and column permutation functions.

As discussed above, at runtime, the constrained interleaver can be efficiently implemented using table lookups, using arrays of pointers and register indirect addressing and/or memory indirect addressing. The FIG. 13 can be used to identify the constrained interleaver's permutation function at design time. Forever after, the identified constrained interleaver's overall permutation and inverse permutation functions can then be implemented using respective passes of incrementing through a respective length-rρn vector of pointers to directly and efficiently at runtime.

To better understand the performance of SC-IRCC with constrained interleaving, it is helpful to consider the weight enumerating function (WEF) of the (n, k) outer block code in the form

$\begin{matrix} {{A\left( {W,L} \right)} = {1 + {\sum\limits_{i = d_{0}}^{n}{\sum\limits_{u = 1}^{k}{a_{i,u}W^{u}L^{i}}}}}} & (40) \end{matrix}$ which can also be written by only considering the weights of the codewords as

$\begin{matrix} {{A(L)} = {1 + {\sum\limits_{i = d_{0}}^{n}{c_{i}L^{i}}}}} & (41) \end{matrix}$ where,

$c_{i} = {\sum\limits_{u = 1}^{k}{a_{i,u}.}}$ The same inner recursive convolutional code that was previously discussed with uniform interleaving is considered for the inner code with constrained interleaving.

Next consider the case when the number of non-zero codewords of the outer code, s (which equivalent to n₀ as discussed previously), is one, i.e., s=1. With constrained interleaving, when s=1, all “1”s of the interleaver are placed along the same row of the interleaver, and the corresponding weight of the interleaver l satisfies, d₀≦l≦n. Further, according to constrained interleaving, this row is randomly selected among all r rows and the contents of the row are randomized among all nρ columns. Hence, the number of possible constrained interleavers when s=1 is given by

$\begin{matrix} {N_{1} = {r\begin{pmatrix} {n\;\rho} \\ l \end{pmatrix}}} & (42) \end{matrix}$ Note that there are

$\quad\begin{pmatrix} {rm} \\ l \end{pmatrix}$ uniform interleavers when the interleaver weight is l [6].

As can be seen from the above, compared with uniform interleaving, constrained interleaving suffers in terms of number of possible interleavers. In order to reduce the gap between the number of interleavers of the two types of interleaving, equation (42) also suggests that it is desirable to employ as small of a value of r as possible, however, by ensuring that r is large enough to maintain the overall minimum weight of the concatenation at d_(f) ⁰d_(f) ^(i). Despite the reduction in the number of interleavers, constrained interleaving eliminates all possible error events when s=1 except for the error event that occurs at the end of termination bits. It can be observed that the minimum weight of the concatenation when s=1 results when l=d₀ and when all of the d₀ “1”s of the interleaver are placed at the lower right corner of the r by rρn interleaver array as highlighted in FIG. 2, and any other arrangement of d₀ “1”s can generate a very high distance of the concatenation. With the proper selection of r, this minimum weight is at least d_(f) ⁰d_(f) ^(i). Denoting the maximum message weight of a minimum weight codeword of the outer code by w_(m), and noticing that c_(d) ₀ in equation (41) is the number of codewords of the outer code with minimum weight d₀, the contribution to error probability by the highlighted bits in FIG. 2 can be bounded as

$\begin{matrix} {P_{e\; 1} < {\frac{w_{m}\rho\; c_{d_{0}}}{k\begin{pmatrix} m \\ d_{0} \end{pmatrix}}{{Q\left( \sqrt{\frac{2d_{0}d_{f}^{i}{RE}_{b}}{N_{0}}} \right)}.}}} & (43) \end{matrix}$

It is seen from equation (43) that in addition to maintaining the weight at the highest possible minimum distance of the concatenation, P_(e1) also achieves interleaver gain. It is noted that the s=1 case with uniform interleaving can have α_(M) and h(α_(M)) in equations (38) and (39) with maximum possible n_(i), and it can also have all other lower values of n_(i) down to n_(i)=1 which are likely to have lower weights for the concatenation.

Next consider the general case of s (1<s≦rρ) non-zero codewords of the outer block code feeding coded bits into the interleaver. In order to focus on the worst case performance contributions, let us consider the case where each of these s codewords has the minimum weight d₀, generating a weight of sd₀ in the interleaver. The corresponding number of possible constrained interleavers can be found by considering the random distribution of s codewords among the rows and considering the randomization of the contents of the rows individually. Let us represent any k th distribution of the codewords among rows in the form of a sequence y _(k)=(y ₁ ,y ₂ , . . . y _(r)), 0≦y _(j)≦Min(ρ,s)  (44) where, y_(j) represents the number of codewords placed in the j^(th) row with

${\sum\limits_{j = 1}^{r}y_{j}} = {s.}$ Denoting the number of non-zero elements of y_(k) by t_(k), the number of possible constrained interleavers resulting from s nonzero outer codewords each with weight d₀ can be written as

$\begin{matrix} {N_{s} = {\sum\limits_{k}{\begin{pmatrix} r \\ t_{k} \end{pmatrix}{\prod\limits_{j = 1}^{r}\;{\begin{pmatrix} {n\;\rho} \\ {y_{j}d_{0}} \end{pmatrix}.}}}}} & (45) \end{matrix}$ For example,

$\begin{matrix} {{N_{1} = {r\begin{pmatrix} {n\;\rho} \\ d_{0} \end{pmatrix}}},{N_{2} = {{\begin{pmatrix} r \\ 2 \end{pmatrix}\begin{pmatrix} {n\;\rho} \\ d_{0} \end{pmatrix}\begin{pmatrix} {n\;\rho} \\ d_{0} \end{pmatrix}{u\left( {r - 2} \right)}} + {{r\begin{pmatrix} {n\;\rho} \\ {2\; d_{0}} \end{pmatrix}}{u\left( {\rho - 2} \right)}}}}} & (46) \end{matrix}$ are the numbers of constrained interleavers that result from one and two non-zero codewords of the outer code respectively, where u(.) is the unit step function. Focusing on the dependence on r, ρ, d₀ and n, it can be seen from equations (45) and (46) that N, is in the order of r^(s)(nρ)^(sd) ⁰ .

The contribution to the error rate in equation (35) made by s non-zero codewords of the outer code each with minimum weight d₀. Since the minimum input weight of an error event of the inner code is two, the maximum number of error events in the inner code is n_(i,max)=d₀└s/2┘, where └.┘ denotes the floor function. Further, when s≧2, the minimum number of error events of the inner code with constrained interleaving (without termination) is d₀. Next consider n_(i) (d₀≦n_(i)≦n_(i,max)) error events of the inner code each with minimum weight h_(m)(j) corresponding to the input weight j, and denote the number of error events with input weight j by x_(j), j=2, 3, . . . , s. The values of x_(j) can be represented in the form of an error event distribution sequence as x=(0, x₂, . . . x_(s)). Note that (a) any x_(j), and hence n_(i) too, is either zero or an integer multiple of d₀ with a maximum possible value equal to

${d_{0}\left\lfloor {s/2} \right\rfloor},{{(b)\mspace{14mu}{\sum\limits_{j = 2}^{s}\; x_{j}}} = n_{i}},{{{{and}(c)}\mspace{14mu}{\sum\limits_{j = 2}^{s}\;{j\; x_{j}}}} = {{sd}_{0}.}}$ Let p=n_(i)/d₀, then the maximum value of p, p_(ma)=└s/2┘. In order to find the contribution from s non-zero codewords of the outer code in equation (35), it is also necessary to find the number of ways n_(i) error events with the associated error event distribution x can be arranged in the interleaver. For any given x, all d₀x_(j) error events are determined by the placement of x_(j) codewords each with weight d₀. Hence, the number of ways n_(i) error events with error event distribution x can be placed in the interleaver is N_(p). Observing that the resulting weight of the coded sequence of the concatenation corresponding to these n_(i) error events of the inner code is

$\begin{matrix} {{d_{a} = {\sum\limits_{j = 2}^{s}\;{x_{j}{h_{m}(j)}}}},} & (47) \end{matrix}$ the corresponding contribution to P_(be) in equation (35) can be written as

$\begin{matrix} {{P_{e\; 2}\left( {s,x} \right)} < {\frac{{sw}_{m}{c_{d_{0}}^{s}\begin{pmatrix} {r\;\rho} \\ s \end{pmatrix}}N_{p}}{{rk}\;\rho\; N_{s}}{{Q\left( \sqrt{\frac{2\; d_{a}{RE}_{b}}{N_{0}}} \right)}.}}} & (48) \end{matrix}$

Equations (47) and (48) can be used to find the significant contributions from all error events that result from s(>1) non-zero codewords of the outer code excluding the error events that occur at the termination. When s>1, the contributions from error events due to termination have a higher distance and a higher interleaver gain than those in equation (43) when s=1, and hence, the contributions made by the error events when s>1, due to termination are negligible.

Note from equation (48) that, for given s, the lowest interleaver gain is achieved by the combination with p=p_(max). Focusing on the dependence on ρ, r, and n, the order of the corresponding error coefficient with the lowest interleaver gain, O(E_(coeff,cons)), is

$\begin{matrix} {{O\left( E_{{coeff},{cons}} \right)} = \left\{ \begin{matrix} {{\frac{{sw}_{m}c_{d_{0}}^{s}}{k}{sw}_{m}c_{d_{0}}^{s}r},^{(\frac{s - 2}{2})}{n^{- \frac{{sd}_{0}}{2}}\rho^{- {({{\frac{{sd}_{0}}{2}s} + 1})}}},{s\mspace{14mu}{even}}} \\ {{\frac{{sw}_{m}c_{d_{0}}^{s}}{k}{sw}_{m}c_{d_{0}}^{s}r},^{(\frac{s - 3}{2})}{n^{- {\lbrack\frac{{({s + 1})}d_{0}}{2}\rbrack}}\rho^{- {({\frac{d_{0}s}{2} + \frac{d_{0}}{2} - s + 1})}}},{s\mspace{14mu}{odd}}} \end{matrix} \right.} & (49) \end{matrix}$ From the dependence on ρ in equation (49), it is observed that the error rate variation in equation (48) achieves interleaver gain for all values of s when d₀≧2. Hence, as with uniform interleaving, all error events with constrained interleaving with an inner recursive code achieve interleaver gain. In addition, it is also seen from equation (49) that it is desirable to use component codes for which

${n^{\frac{d_{0}}{2}} > {rc}_{d_{0}}},$ as this can decrease the error coefficient with increasing values of s. However, the latter condition may not be that important for many combinations of component codes due to the increase in the weight of the concatenation with increasing values of s.

One important contribution in equation (48) is the one that corresponds to the minimum weight of the concatenation, which with constrained interleaving is maintained at d₀d_(f) ^(i). Note that the minimum weight of the inner code is

${d_{f}^{i} = {{\begin{matrix} {Min} \\ u \end{matrix}{h_{m}(u)}} = {h_{m}(\lambda)}}},$ where λ is the input weight of the inner code that generates the minimum weight of the code. The minimum weight of the concatenation results from s=λ, non-zero outer codewords of the outer code each with weight d₀ when p=1. Hence, the contribution to P_(be) corresponding to the minimum weight of the concatenation is given by

$\begin{matrix} {P_{e\; 3} < {\frac{\lambda\; w_{m}{c_{d_{0}}^{\lambda}\begin{pmatrix} {r\;\rho} \\ \lambda \end{pmatrix}}N_{1}}{{rk}\;\rho\; N_{\lambda}}{{Q\left( \sqrt{\frac{2\; d_{0}d_{f}^{i}{RE}_{b}}{N_{0}}} \right)}.}}} & (50) \end{matrix}$ It is seen the error coefficient of the variation in equation (50) can decrease fast with increasing ρ especially at higher values of λ.

The contributions in equations (43) and (49), the P_(be) variation with constrained interleaving can be written as

$\begin{matrix} {P_{e,{constrained}} < {P_{e\; 1} + {\sum\limits_{{s \geq 2},x}\;{{P_{e\; 2}\left( {s,x} \right)}.}}}} & (51) \end{matrix}$ It is noted that depending on the component codes, the interleaver size and the SNR, the error rate can be dominated by one of the variations in equation (51). It is likely that at very low error rates the variation with the lowest distance given by P_(e3) in equation (50) dominates the overall performance. Similarly, at lower SNR values it is likely that the variation with the lowest interleaver gain (that is likely to be the term in equation (48) with s=2 and p=p_(max)=1) dominates the overall performance.

The number of rows of the interleaver, r, is selected to ensure that the overall minimum distance is strictly maintained at d₀d_(f) ^(i). However, for a given interleaver size N=rρn, by sacrificing the minimum distance, it is possible to increase ρ thereby increasing the interleaver gain. Hence, even though the selection of r, to guarantee the minimum distance at d₀d_(f) ^(i) is a good starting value of r, depending on the desired error rates and component codes, it may be possible to improve performance by lowering the value of r, and sacrificing the minimum distance slightly. The final best value of r, can be numerically found using the bound in equation (51) depending on the application.

We next compare the performance of SC-IRCC with constrained interleaving to SC-IRCC implemented with uniform interleaving. It is recommended in the literature [6] that serial concatenation be used with odd values of d₀ with uniform interleaving. Hence, we compare SC-IRCC that uses constrained interleaving with uniform interleaving when d₀ is odd. In order to carry out the comparison, it is first necessary to develop the error rate variation of uniform interleaving with an outer block code and an inner recursive code. Even though in uniform interleaving it is not necessary to consider the interleaver in a row/column format, for comparison with constrained interleaving, we consider the same row/column format for the uniform interleaver too which is equivalent to a uniform interleaver with size N=rρn.

Consider s≧1 non-zero codewords of the outer code. Since uniform interleaving has no structure to control error events, in order to capture the significant contributions in equation (35), we consider all possible weights (not just weight d₀) of the s codewords. Let us consider s codewords with weights expressed in a sequence as u=(u₁, u₂, . . . , u_(s)); d₀≦u_(j)≦n. We can group these codewords into e(≧s) non-empty groups, g₁, g₂, . . . , g_(e) according to their weights so that weights of all codewords in group g_(j) is the same which is denoted by d(g_(j)) and the number of codewords in group g_(j) is z(g_(j))(≧1). Note that

${\sum\limits_{j = 1}^{e}\;{z\left( g_{j} \right)}} = {s.}$ The corresponding interleaver weight is

$l = {{\sum\limits_{j = 1}^{s}\; u_{j}} = {\sum\limits_{j = 1}^{e}\;{{d\left( g_{j} \right)}{{z\left( g_{j} \right)}.}}}}$ The maximum number of error events of the inner code is n_(i,max)=└l/2┘, while the minimum number of error events is one in contrast to the minimum number of d₀ error events in constrained interleaving. Consider the case of n_(i) (1≦n_(i)≦n_(i,max)) error events of the inner code each generating the minimum weight of the coded bits for that input weight, and denote the input weight of the j th error event by q_(j). These input weights can be expressed in an error event distribution sequence as q=(q₁, q₂, . . . , q_(n) _(i) ), 0≦q_(j)≦n_(i). Note that

${\sum\limits_{j = 1}^{n_{i}}\; q_{j}} = l$ and the weight of the coded sequence of the concatenation is

$\begin{matrix} {d_{uni} = {\sum\limits_{j = 1}^{n_{i}}\;{{h_{m}\left( q_{j} \right)}.}}} & (52) \end{matrix}$ Hence, the contribution to P_(be) made by s non-zero codewords of the outer code with error event distribution q is

$\begin{matrix} {P_{ed} < {\sum\limits_{{s \geq 1},u,q}\;{\frac{\left( {\sum\limits_{k = 1}^{s}\; w_{u_{k}}} \right)\begin{pmatrix} {r\;\rho} \\ s \end{pmatrix}\left( {\prod\limits_{k = 1}^{s}\; c_{u_{k}}} \right)\begin{pmatrix} {{rn}\;\rho} \\ n_{i} \end{pmatrix}{s!}}{k\;\rho\;{r\begin{pmatrix} {{rn}\;\rho} \\ l \end{pmatrix}}\left( {\prod\limits_{j = 1}^{e}\;{\left\lbrack {z\left( e_{j} \right)} \right\rbrack!}} \right)}{{Q\left( \sqrt{\frac{2\; d_{uni}{RE}_{b}}{N_{0}}} \right)}.}}}} & (53) \end{matrix}$

The performance with uniform interleaving can be found by using equation (53) and summing over significant combinations of s, u and q. It can be seen that there are significant contributions with s=1. Recall that all regular merging events with s=1 are eliminated with constrained interleaving. In addition to generating smaller weights of the concatenation, the s=1 case can achieve the smallest possible interleaver gain too. It follows from equation (53) (and from equation (37)) that the minimum interleaver gain for given l is achieved with n_(i)=n_(i,max). Hence, when d₀ is odd, the weight d_(f,eff) ^(i) (d₀+1)/2 for the concatenation is generated from a single codeword (s=1) with weight (d₀+1) of the outer code. It follows from equation (47) that in constrained interleaving a similar term with a weight of d₀d_(f,eff) ^(i) results from two codewords (s=2) of the outer code. Clearly, the weight with constrained interleaving is higher than that with uniform interleaving. Further, even though for given values of s and l, the number of uniform interleavers is higher, the comparable terms of the two interleavers usually result from two different values of s, and hence, the actual comparison of the error coefficients can also favor constrained interleaving over uniform interleaving. For example, considering the dependence on r, ρ and n corresponding to the above two weights, it follows from equations (48) and (53) that the ratio of the error coefficients of constrained interleaving to uniform interleaving is on the order of [r^((d) ⁰ ^(+1)/2)n^(−(d) ⁰ ^(−1)/2)ρ^(−(d) ⁰ ^(−3)/2)]. Hence, for smaller values of r, (compared with the product nρ) constrained interleaving can have smaller error coefficients in addition to the higher distances of the corresponding terms.

The focus in the design of serial concatenation with uniform interleaving is to achieve the maximum interleaver gain and not to remove terms that correspond to lower weights in equation (35). With that focus some of the lower weight terms can also end up achieving the lowest interleaver gain as can be seen from equation (38) for the case corresponding to s=1, l=(d₀+1) and n_(i)=l/2 when d₀ is odd. Hence, these lower weight terms with minimum interleaver gain can dominate the overall error rate in equation (35). Constrained interleaving on the other hand removes lower weight terms and also achieves interleaver gain in the remaining terms. Even though the interleaver gains of the two types of interleavers at any given l, compare favorably for uniform interleaving, the interleaver gains of constrained interleaving at similar type of weights can be lower than those of uniform interleaving. Hence, constrained interleaving can perform better than uniform interleaving at the same interleaver size or can be used to improve the performance over that of uniform interleaving with smaller interleaver sizes.

Another inherent undesirable property of uniform interleaving is the existence of its error rate floor which can be an important consideration especially at low error rate applications such as in optical communications and in magnetic recording. The reason for the relatively high error rate floor is due to the presence of low weight codewords of the concatenation. Specifically, the minimum weight of the concatenation h_(m) is the minimum of all h_(m)(l), or any combinations of h_(m)(l_(i))s with

${\sum\limits_{i}\; l_{i}} = l$ that correspond to a valid weight of the interleaver l generated by the outer code. On the other hand, constrained interleaving achieves the highest possible minimum weight of the concatenation that has the corresponding error rate variation in equation (43). Hence, the performance gain of constrained interleaving over uniform interleaving can be even more significant at low error rates.

In addition to achieving performance gains, constrained interleaving also has other advantages over uniform interleaving due to a smaller interleaver size. The smaller interleaver size of constrained interleaving reduces the delay and the memory requirement of the decoder. It also reduces the computational complexity by reducing the number of iterations when iterative decoding is used. In order to minimize the number of iterations, it is desirable to employ a stopping criterion, among many that have been discussed in the literature to stop the iterations. These various stopping criteria decide to stop iterations based on the invariability of the decoded bits within a frame. The invariability of decoded bits is measured using various respective metrics. Since it is more likely to find variations in the decoded bits within a frame when the frame size is larger, the average number of iterations with a longer interleaver is higher than that with a shorter interleaver at the same error rates. This is supported by the numerical results reported in the literature. Hence, decoding with a constrained interleaver that has a smaller interleaver size, on average, requires a fewer number of iterations than decoding with a much larger uniform interleaver. Since the number of computations per bit in a single iteration is the same for decoding with both interleavers, the total decoding computational complexity with constrained interleaving is therefore lower than that with uniform interleaving. The exact amount of saving in complexity depends on the component codes, the sizes of the two interleavers, and the operating error rate.

The recovery of channel state information (CSI) can also be simpler with constrained interleaving. If the channel is a slow varying channel, a decision feedback equalizer (DFE) that neglects any variations of the channel within a frame and uses the decoded bits to estimate the channel for the next frame can be better constructed with a decoder that has a smaller interleaver size than with uniform interleaving. With a smaller interleaver size, the channel is more likely to remain constant over a frame, and the estimated channel parameters by the DFE are more likely to be the channel parameters for the next frame. A similar advantage can be found if joint channel estimation and decoding is employed. It is known that joint channel estimation and decoding is possible with iterative decoding by updating channel information along with extrinsic information during iterations. However, such joint channel estimators/decoders require a significantly large number of iterations. If joint channel estimation and decoding is used, compared with uniform interleaving, constrained interleaving with a smaller interleaver will require a lower number of iterations as it can stabilize the channel estimates and the decoded bits faster, thereby reducing the complexity. The difference in number of iterations between constrained and uniform interleaving is likely to be higher with joint channel estimation and decoding than with decoding only. However, if CSI is recovered using training sequences which can be done prior to decoding, the CSI recovery will be independent of the type of the interleaver.

So far we have been considering block codes for the outer code of the concatenation. We next discuss how trellis based convolutional codes can also be used as the outer code along with an inner recursive convolutional code. It is known that the outer code can be either recursive or non-recursive, and further non-recursive outer convolutional codes perform slightly better than recursive outer convolutional codes [6].

In case of outer convolutional codes, regardless of the type of the interleaver, the outer code should be terminated at the end of every block. In case of constrained interleaving, the coded bits of the outer code with the termination are used to fill up the r by m rectangular array. It is noted that, unlike selecting the value of m as a multiple of the codeword length as in case of outer block codes (ρn), the value of m can be arbitrarily selected in case of outer convolutional codes. When extending constrained interleaving with outer block codes to outer convolutional codes, care should be taken due to the fact that error events can start from any bit, where as in case of block codes these errors are restricted to codewords of length n which have well defined starting and ending points. In order to accommodate for this change, it is necessary to modify the constrained interleaving procedure from that of outer block codes described in connection with FIG. 13. Specifically, constrained interleaving with outer convolutional codes can be implemented according to the following three actions:

-   -   1. Feed the coded bits of the outer code into the r by nρ array         along rows starting from the first row.     -   2. Randomize the contents of the rows independently. This action         should be modified from that of outer block codes. It can be         done in two different methods, Method 1 and Method 2, depending         on the selected scheme as discussed below.     -   3. Shuffle the r rows without changing the contents in them.     -    *Shuffled_Rows=Rand(*Rows),     -    where *Rows is the r—element vector of pointers to the rows of         the constrained interleaver and *Shuffled_Rows is a vector of         pointers to the randomized-ordered rows of the constrained         interleaver after row shuffling.

Action 2 above can be implemented in two different ways depending on the construction of the concatenated code. One easy method to maintain the same randomization method used with outer block codes (810) is to remove the influence of the last bits of any row from the starting bits of the next row. This can be done by terminating every row separately which is referred to as Method 1 here. In this method action 2 above will be identical to action 2 with outer block codes 810.

It is also possible to use a different method without terminating rows individually which leads to Method 2. Method 2 focuses on separating the fast several bits, say n bits, of any row from the first n bits of the next row to overcome their dependence without terminating every row separately. The value of n can be chosen to be the path memory length of the outer code which is the length of any non-zero coded sequence that has a weight of at least the minimum distance of the code. The last n bits and the first n bits of two different rows can be separated by first selecting a set of m_(mid) columns placed in the middle of the r by nρ array and preserving the right hand side and the left hand side of it for the last n bits and the first n bits respectively during 810. In other words, during the 810, the last n bits of any row are randomized only over the columns right of the m_(mid) identified columns. Similarly, the first n bits of every column are randomized only over the columns left of the m_(mid) columns. However, other bits of any row are randomized over all columns. The value of m_(mid), that depends on the inner code, should be selected to maintain the minimum distance of the concatenation at d₀d_(f) ^(i). Even though this additional restriction on the first n and the last n bits of every row reduce the number of possible constrained interleavers, its impact diminishes with increasing values of m.

To help understand the performance of SC-IRCCs implemented with constrained interleaving, we present numerical results comparing constrained interleaving with uniform interleaving. Consider a (7,4) outer Hamming code with d₀=3 along with a 4-state recursive inner code with generating matrix

$\begin{matrix} {{G(D)} = {\left\lbrack {1\mspace{14mu}\frac{1 + D^{2}}{1 + D + D^{2}}} \right\rbrack.}} & (54) \end{matrix}$ The code in equation (54) has d_(f) ^(i)=5, d_(f,eff) ^(i)=6 and h_(m)(3)=5. Analyzing the error events of the above inner code, it can be found that r=4 is sufficient to maintain the minimum distance of the concatenation with constrained interleaving at d₀d_(f) ^(i)=10 among all error events except for the error event that corresponds to the termination highlighted in FIG. 12 that has the error rate variation in equation (43). Depending on the application and the size of the interleaver, if the variation in equation (43) is negligible r=4 can be used, and if not r=8 should be used that guarantees the overall minimum distance at 15.

FIG. 14 shows the error rate variations of uniform and constrained interleaving of an SC-IRCC using an outer (7,4) Hamming code and a rate 1/2 inner recursive convolutional code when the interleaver lengths of are set to N=112 and N=336 and N=1008. Performance curves for three different interleaver sizes are shown. As can be seen, much lower error rates are reached by the concatenated code with the constrained interleaver than with a uniform interleaver of the same length. This allows much shorter interleavers to be used to reach a target bit error rate for a given signal to noise ratio in a practical implementation.

FIG. 15 shows the error rate variations of constrained interleaving with a (7,6) outer SPC code and along with a rate 1/2 inner convolutional code of equation (54). FIG. 15 compares SC-IRCC implemented with uniform interleaving and with constrained interleaving when the interleaver lengths of are set to N=112 and N=336 and N=1008. In the literature full rate recursive inner codes have been used to improve the overall rate. It is seen from FIG. 15 that constrained interleaving performs better than uniform interleaving and constrained interleaving achieves interleaver gains that are similar or better than those with uniform interleaving. It is also seen that constrained interleaving can achieve better performance with a much smaller interleaver size, and the improvement becomes more significant at lower error rates.

Other embodiments use (8,7) and (4,3) SPC outer codes (k=7 and k=3) with rate 1/2 IRCCs with v=2 and v=3. This can be modulated by BPSK, 4-PAM or some other modulation format like QAM and transmitted over a channel. The constrained interleaving based structures as presented in FIGS. 4-6 for SC-IRCC are then applied to reduce the number of needed decoder iterations. The number of columns of the interleaver is left as an adjustable parameter than can be optimized for a given code, modulation type and channel and a value that optimizes performance is preferably selected. In order to ensure the maximum possible minimum distance, 8 rows when v=2 and 11 rows when v=3 can be used. The impact of reducing the number of rows to a point that sacrifices the minimum distance constraint can also be considered if the reduction in error coefficient offsets and surpasses the lost due to distance.

So far with constrained interleaving with IRCC targets an overall minimum distance of d₀d_(f) ^(i). Due to the nature of recursive codes, by sacrificing interleaver gain, the minimum distance of the code can be further improved. For example, let us consider the serial concatenation of the (7,6) outer code with the IRCC given by (54) with constrained interleaving considered in FIG. 15. The error event with minimum distance with probability (50) is resulted by having two non-zero columns with (1 1 1) segments generated from three weight two SPC codewords placed on three rows. This error event can be prevented by further constraining the interleaver to not allow more than a single bit (in general (d₀−1) bits) from two different outer codewords to be placed along any single column of the array. This not only prevents the error events with d₀d_(f) ^(i), it also prevents all error events with distance d₀h_(m)(j) of the concatenation for j≧2. Hence, with this extended constraint the minimum distance of the overall concatenation with the added constraints can be increased beyond d₀d_(f) ^(i) and the value it reaches can be controlled by the additional constraints put on the interleaver. The number of rows can be appropriately increased depending on the target minimum distance, or alternatively, additional constraints can be imposed to place coded bits of the same codeword of the outer code along the same row with a minimum separation of preselected number of rows between any two bits of that codeword.

The constrained interleaver with additional constraints can be implemented by first constructing the interleaver as described in FIG. 13 and checking for additional constraints. If all additional constraints are satisfied the interleaver is selected for application, and if not, additional work is required. The additional work can be listed as:

1. Randomly select a row from the interleaver found by FIG. 13 as that row of the new interleaver. Check for constraints. If all are satisfied move to step 2. If not, keep randomizing that row until all constraints are satisfied. (There are constraints in this step only if a minimum column separation is required between coded bits of the same codeword)

2. Randomly select another row. Check for all individual and joint constraints with already selected rows. If all constraints are satisfied move to step 3. If not, keep randomizing the selected row again until all constraints are satisfied.

3. Repeat step 2 until all rows are selected.

The extended constraints reduce the available number of interleavers thereby reducing the interleaver gain. Hence, these additional constraints in constrained interleaving provide a tradeoff between the distance and the interleaver gain. The best tradeoff can be selected based upon numerical simulation studies that look for the best set of constraints to be used for a particular set of codes and/or modulation/mapping schemes, depending on the application.

Any or all of the constrained interleaving tecyhniques as discussed herein can also be applied to parallel concatenation (such as turbo codes). However, this can only guarantee that the second constituent code can spread the error events. As a result, it cannot guarantee the product of the distances for the concatenation. Due to the improvement in the second constituent code, the constrained interleaving methods, apparatus, and systems presented herein can improve performance of parallel concatenated codes over uniform interleaving. In the case of the parallel concatenated codes, the additional constraints described in the above can also be used. This provides a means to improve interleavers such as those disclosed in U.S. Pat. No. 6,857,087 due to a higher interleaver gain and due to having a target overall minimum distance to control the design.

In the 4-PAM (16-QAM) case, this embodiment can be used to improve upon the rate 1/2 CTC that has been adopted in the WiMAX standard with an interleaver size of 960. For example, even though the above serially concatenated cases have slightly a lower rate (specifically, rate of the concatenation is k/[2(k+1)]) than the CTC, they perform significantly better with a shorter interleaver length and a lower decoding complexity.

It should be noted that the SC-IRCC approach with constrained interleaving is an attractive alternative to communication standards that use turbo codes such as 3GPP and 3GPP2. For example, much shorter interleavers and simpler codes can be used to achieve the same bit error rate performance. The BICM schemes used in 802.11a/g and 802.16 can also be replaced with a more efficient SC-IRCC coding scheme that makes use of constrained interleaving. All such system level embodiments are contemplated by the present invention. It is also contemplated that SC-BC and SC-IRCC can be used in the encoding of backbone optical links and for magnetic recording channels.

As discussed previously, the transmitter 300 can be implemented to generate improved trellis coded modulation schemes by selecting the inner code to be a non-recursive convolutional code (trellis code). When the mapper 320 is used, we call this improved form of trellis coded modulation SCITC (serial concatenation with inner trellis code).

For demonstration, we consider a SCITC scheme that employs an outer (n, k) block code with minimum Hamming distance d₀ and an inner trellis code constructed by memory v convolutional code followed by a mapper as illustrated in FIG. 3. Denoting the raw MSED value of the inner trellis code corresponding to an interleaver weight u by D_(u) ², the overall MSED of the concatenation with uniform interleaving can be written as

$\begin{matrix} {D_{\min,{uni}}^{2} = {{\begin{matrix} {Min} \\ {u \geq d_{0}} \end{matrix}D_{u}^{2}} = D_{u_{m}}^{2}}} & (55) \end{matrix}$ where, u_(m) is the value of u that minimizes D_(u) ² in equation (55). It is seen from equation (55) that the impact of d₀ on the MSED is simply to prolong the error event that determines the minimum distance, and hence, its impact on the MSED is not usually that significant. In constrained interleaving with SCITC, the objective is to achieve the highest possible MSED for the concatenation while preserving the advantages of interleaving. The constrained interleaver is constructed using the method shown FIG. 13 and the related discussion. That is, the constrained interleaver for SCITC is implemented as the IRCC case described above. When the constrained interleaver 1300 is used, any codeword of the outer code have the freedom to be placed in any row, codewords have the freedom to get mixed up randomly, and coded bits of any codeword get placed along the same row of the interleaver.

When the value of r is large enough to ensure that the all non-zero coded bits of the outer code are pushed into different error events of the inner code with minimum distance, the concatenation can achieve the highest achievable MSED. Specifically, the constrained interleaving can achieve an overall MSED of

$\begin{matrix} {D_{\min,{cons}}^{2} = {{d_{0}\begin{matrix} {Min} \\ u \end{matrix}D_{u}^{2}} = {d_{0}D_{u_{\min}}^{2}}}} & (56) \end{matrix}$ for the concatenated code, where u_(min) is the input weight that minimizes the MSED of the inner code. Due to the linear dependence of the MSED on d₀ in equation (56), the

MSED with constrained interleaving can be significantly higher than that with uniform interleaving.

Consider the case where the SCITC is transmitted over an AWGN (additive white Gaussian noise) channel with power spectral density N₀/2. Our desire is to compare SCITC with constrained interleaving to the same SCITC, but implemented with uniform interleaving. We use the union bound approach as in (35) for the analysis and consider the contributions to the bit error rate, P_(be), from the dominant terms in the bound. Throughout this analysis, the distance and the weight of the outer code refers to the Hamming distance and the Hamming weight while the distance of the inner code or of the concatenation refers to the Euclidean distance. For the analysis, we again consider the weight enumerating function (WEF) of the (n, k) outer block code in the form [6]:

$\begin{matrix} {{A(L)} = {1 + {\sum\limits_{i = d_{0}}^{n}\;{c_{i}L^{i}}}}} & (57) \end{matrix}$ where, c_(i) is the number of codewords with weight i.

Consider the impact of a single non-zero codeword of the outer code with minimum weight d₀. With constrained interleaving, all of the d₀ “1”s in the interleaver will be placed along a single row. Realizing that there are r ways to select a row and

$\quad\begin{pmatrix} {n\;\rho} \\ d_{0} \end{pmatrix}$ ways to select columns on that row, the corresponding number of constrained interleavers can be written as

$\begin{matrix} {N_{1} = {{r\begin{pmatrix} {n\;\rho} \\ d_{0} \end{pmatrix}}.}} & (58) \end{matrix}$ In the inner code each of these “1”s can generate a separate error event with MSED D₁ ², making the total MSED of the concatenation d₀D₁ ². Since there are rρ ways to select a single non-zero codeword of the outer code, and N₁ ways to have d₀ error events in the inner code, the corresponding contribution to P_(be) resulting from a single non-zero codeword of the outer code with weight d₀ can be bounded as

$\begin{matrix} {P_{{e\; 1},{cons}} \leq {\frac{w_{d_{0}}c_{d_{0}}}{k}{Q\left( \sqrt{\frac{d_{o}D_{1}^{2}}{N_{0}}} \right)}}} & (59) \end{matrix}$ where, w_(j) denotes the maximum message weight of a codeword with weight j of the outer code. The inequality in equation (59) results from the fact that the message weight for some codewords with weight d₀ can be smaller than w_(d). It is seen from equation (59) that P_(e1,cons) does not achieve any interleaver gain.

Next compare equation (59) with the error rate variation of the corresponding case with uniform interleaving. Even though in uniform interleaving it is not necessary to consider the interleaver in a row/column format, for comparison with constrained interleaving, we consider the same row/column format for the uniform interleaver too which is equivalent to a uniform interleaver with size N=rnρ. When the interleaver weight is d₀, the inner code can have any l number of error events, where 1≦l≦d₀. The error rate variation when l=1 can be found by realizing that are

$\quad\begin{pmatrix} {{rn}\;\rho} \\ d_{0} \end{pmatrix}$ uniform interleavers, and there are rnρ ways to have a single merging event in the inner code (as in the literature the length of the error events are neglected here). Hence, the corresponding contribution to P_(be) is

$\begin{matrix} {P_{{e\; 1},1,{uni}} \leq {\frac{w_{d_{0}}c_{d_{0}}r}{k\begin{pmatrix} {{rn}\;\rho} \\ d_{0} \end{pmatrix}}{Q\left( \sqrt{\frac{D_{d_{0}}^{2}}{N_{0}}} \right)}}} & (60) \end{matrix}$

Clearly, the variation in equation (60), has a lower distance but achieves interleaver gain for d₀≧2 as the error coefficient can be lowered by increasing ρ. Similarly, when l=d₀, the contribution to P_(be) is identical to equation (59). Hence, it is seen that uniform interleaving has lower weight terms that can achieve interleaver gain, and their effect can be made insignificant by increasing the size of the interleaver. However, the error rate with uniform interleaving cannot be lowered below that in equation (59). Hence, the performance with uniform interleaving is lower bounded by equation (59). It is further mentioned that the impact of multiple number of non-zero codewords can increase the distance but can have error coefficients that increase with increasing interleaver size. For example, when s non-zero codewords, each with weight d₀, generate sd₀ error events in the inner code, its contribution to P_(be) with constrained and uniform interleaving are both given by

$\begin{matrix} {P_{{e\; 2},{uni}} \leq {\frac{{sw}_{d_{0}}{c_{d_{0}}^{s}\begin{pmatrix} {r\;\rho} \\ s \end{pmatrix}}}{r\;\rho\; k}{{Q\left( \sqrt{\frac{{sd}_{o}D_{1}^{2}}{N_{0}}} \right)}.}}} & (61) \end{matrix}$ Clearly, the error coefficient in equation (61) increases with increasing ρ for s≧2. Hence, in uniform interleaving, when the interleaver size is increased to reduce the impact of the lower weight terms, the contribution from these higher weight terms can become significant particularly at low to medium signal to noise ratio (SNR) values. Hence, constrained interleaving can achieve the best achievable performance with uniform interleaving with much smaller interleaver sizes. Using analysis similar to the SC-BC and SC-IRCC cases described above, it can also be shown that constrained interleaving also has error coefficients that increase with interleaver size. However, since constrained interleaving can perform well at smaller interleaver sizes the impact of contributions that have increasing error coefficients with interleaver size can be maintained at insignificant levels.

A similar analysis as performed hereinabove for the SC-IRCC case can be performed for the SC-IRCC case where the distance measure of the inner code is the Euclidian distance instead of the Hamming distance. Similar gains are achieved and it is recalled that modified trellis coded modulation schemes can be constructed using an IRCC and the transmitter 400. Such schemes are known to perform better than versions that use non-recursive convolutional codes and a mapper as the inner code. Our simulation studies have confirmed the analysis but the details of this analysis and the simulation results are not presented here due to their repetitive nature relative to what has already been presented hereinabove in the many other examples and cases.

With that in mind, we have considered the serial concatenation of a (10,9) outer SPC code that has d₀=2 with an inner (64,45) extended BCH code that has d_(i)=8 (see S. Lin and D. Costello, Jr., Error Control Coding: Fundamentals and Applications, 2^(nd) Ed., Pearson Prentice-Hall, 2004.) This combination of component codes generates a concatenated code with rate R=0.6328. FIG. 16 shows the bit error rate performance curves of this SC-BC implemented with constrained and uniform interleaving. Note that constrained interleavers of size N=450, N=900, N=1800 all come much close to the lower bound than the same code using a uniform interleaver of length 4500.

For a base reference and comparison to other codes, the Shannon limit is also plotted in FIG. 16. The Shannon limit has been calculated by using the expression for the capacity C in one-dimensional signaling given by:

$\begin{matrix} {C = {\frac{1}{2}{\log_{2}\left( {1 + \frac{E_{b}}{N_{0}}} \right)}}} & (62) \end{matrix}$ The above equation calculates the minimum required SNR to reduce the error rate below any desired value when C is equal to the rate of the code R. That is, if C=R is plugged into equation (62), then after a simple manipulation, equation (62) can be written as:

$\begin{matrix} {\left( \frac{E_{b}}{N_{0}} \right)_{Shannon} = {10\;{\log\left( {2^{2\; R} - 1} \right)}{({dB}).}}} & (63) \end{matrix}$ Equation (63) gives a direct expression for the Shannon limit in terms of the code's rate. This limit helps one to determine the quality and power of the code and to compare it to other codes using the Shannon limit as a reference.

It is seen that from FIG. 16 that the SC-BC implemented with constrained interleaving performs significantly better than when the same SC-BC is implemented with uniform interleaving. It is seen that the performance of constrained interleaving approaches the lower bound as the interleaver size increases. Note that the size of the interleaver is 450ρ. Also note that the performance of the SC-BC implemented with constrained interleaving more or less meets the Shannon limit at error rates in the 10⁻⁵ region and is within a dB of the Shannon limit even at error rates as low as 10⁻⁸. Further, as expected, the gain of constrained interleaving over uniform interleaving increases as the error rate decreases. Hence, constrained interleaving is very attractive for optical communications (which target error rates around 10⁻¹²) and for magnetic recording (which targets error rates around 10⁻¹⁶) type applications.

An important use of the Shannon limit is to be able to compare the strength and quality of different types of codes. Many communication standards make use of Turbo codes. Some examples of communication standards that use Turbo codes are 3GPP CDMA cellular air interfaces as well as WiMAX OFDMA. For comparison purposes, we make reference to two more articles in the literature: [17] S. Benedetto and G. Montrosi, “Unveiling of turbo codes: Some results on parallel concatenated coding schemes”, IEEE Trans. on Inform Theory, vol. 42, pp. 409-428, March 1996; and [18] U. Wachsmann, R. F. H. Fischer and J. B. Huber, “Multilevel Codes: Theoretical concepts and practical design rules”, IEEE Trans. on Inform Theory, vol. 45, 1361-1391, July 1999.

The performance curves of FIG. 16 show that the performance of serial concatenated codes with constrained interleaving can be closer to the Shannon limit than turbo codes [17] and multi-level codes [18] while maintaining a shorter interleaver. For example, see FIG. 15 of [17] that presents results of a rate 1/3 turbo code for different memory lengths and interleaver sizes. It follows from equation (63) that the Shannon limit when R=1/3 is −2.31 dB. It can be seen from FIG. 15 that while the SC-BC performs effectively at the Shannon limit at an error rate of 10⁻⁵, the best Turbo code (16-state version with length N=1000 interleaver) presented in FIG. 15 of [17] is about 2.5 dB from the Shannon limit at the same error rate of 10⁻⁵. From FIG. 16, and by drawing a horizontal line at the 10⁻⁸ error rate, it is seen that the SC-BC with a constrained interleaver of size N=450 is roughly 2 dB away from the Shannon limit. The best turbo code shown in FIG. 15 of [17] is roughly 2.31+2.5=4.81 dB away from the Shannon limit at the same error rate of 10⁻⁸. Hence, it is seen that an SC-BC with constrained interleaving can be designed to significantly perform better than turbo codes with respect to the Shannon Limit.

Similarly, FIG. 10 of [18] shows that multilevel codes that employ long interleavers (like 20,000 bits) are also about 1 to 1.5 dB away from the Shannon limit at error rates around 10⁻⁵. Hence, it is seen that an SC-BC with constrained interleaving can be designed to significantly perform better than multilevel codes with respect to the Shannon Limit.

As discussed above, SC-IRCCs can achieve interleaver gain well below the lower bound that limits interleaver gain in SC-BC's. Hence, compared to SC-BCs, SC-IRCCs may be implemented with component codes with lower minimum distance and still produce good results. Further, in order to increase the interleaver gain, it is desirable to increase the number of columns m=nρ of the interleaver array. This implies that when designing SC-IRCCs for use with constrained interleaving, it is desirable to use inner codes for which the minimum required number of rows r is low. As an example, consider an SC-IRCC implemented using a (15,10) extended Hamming code with d₀=4 as the outer code, and a rate 2/3 punctured recursive convolutional code with memory v=2 as the inner code. The minimum distance of this concatenation can be maintained at d₀d f with constrained interleaving by employing r=4 rows in the interleaver array. The rate 2/3 inner code is constructed starting with a rate 1/2 recursive convolutional code and using the puncturing pattern (1011). Such high rate punctured convolutional codes are documented in the literature [10,11].

FIG. 17 shows the bit error rate performance curve of a serial concatenation of an outer (15,10) extended Hamming code and an inner code that is a rate 2/3 punctured recursive convolutional code with 4 states. The Shannon limit is also plotted. Again it is noticed that the SC-IRCC with constrained interleaving achieves interleaver gain. Also, this SC-IRCC performs significantly better than when implemented with uniform interleaving. Importantly, this SC-IRCC performs much closer to the Shannon limit at the 10⁻⁵ error rate than Turbo codes and multilevel codes as discussed in [17,18] with a much shorter interleaver.

The best Turbo code reported in [17] uses 16 states and interleaver length of N=1000 to get to within 4.81 dB of the Shannon limit at the 10⁻⁸ error rate. This can be compared to the SC-IRCC of FIG. 17 that uses an inner code with only two states and an interleaver of length N=120 to similarly get to within 4.81 dB of the Shannon limit. Alternatively, if improved performance is desired, this same SC-IRCC with its 2 state inner code can be used with an interleaver of length N=240 to perform within about 2.8 dB of the Shannon limit at the 10⁻⁸ error rate.

At this point some design methods are presented for the design of serial concatenated codes that are targeted for implementation with a constrained interleaver. In the design of SC-BC's, the objective is to try to achieve the performance lower bound of the concatenation. As it is seen from equation (8), the lower bound is determined by the product of the minimum distances d₀d_(i), while the error coefficient λ₂ depends on the number of codewords of the outer code with minimum distance d₀. For two selected block codes, while the product of d₀d_(i) does not depend on which code is selected as the outer or the inner code, the error coefficient λ₂ can be lowered by selecting the code that has the lower number of codewords with minimum distance as the outer code. Usually smaller codes have lower number of codewords with minimum distance. Therefore it is most often desirable to employ the smaller code as the outer code in serial concatenation of block codes with constrained interleaving. In general, the design approach is to select the component codes to provide a desired d₀d_(i) and to minimize the error coefficient λ₂. However, if the interleaver gain is the biggest focus, it is possible to employ the bigger code as the outer code and the smaller code as the inner code.

In addition, the constrained-interleaved SC-BC approach can be applied to non-binary codes like non-binary BCH codes (BCH=BCH code's inventor's initials) and RS (Reed-Solomon) codes. The most desirable way to handle non-binary codes is to do the coding on non-binary symbols and then convert the coded symbols back to binary bits for interleaving and transmission. The transmission can however be done by mapping bits on to higher order symbols through a mapper. As discussed before with block codes, non-binary codes which are usually powerful codes can be preferably used as the inner code. For example, if a powerful RS code is used as an inner code its minimum distance can be doubled by employing an outer SPC code and employing constrained interleaving thereby targeting a 3 dB gain. However, by targeting the interleaver gain the RS code can be used as the outer code and the SPC can be used as the inner code. In this configuration, preferably the codewords of the RS code can be converted back into bits and constrained interleaved. However, if desired, the interleaving can also be done on symbols. Interleaving on bits increases the number of columns and thereby increases the interleaver gain. This class of SC-BCs designed using constrained interleaving have potential applications in high speed communications such as in systems that follow the ITU G.709 standard.

Non-binary codes can also be used with constrained interleaving with inner recursive convolutional codes to generate attractive SC-IRCCs. Some specific design methods can be summarized as follows:

-   -   1. Use a powerful RS outer code with a full rate IRCC. This does         not change the minimum distance of the RS code but due to the         IRCC it can achieve interleaver gain.     -   2. Use a powerful RS outer code with high rate IRCC. This can         increase the minimum distance and achieve interleaver gain. High         rate recursive convolutional codes are found in the literature,         e.g., see [10] or [19] F. Daneshgaran, M. Laddomada and M.         Mondin, “An extensive search for good punctured rate k/(k+1)         recursive convolutional codes for serially concatenated         convolutional codes”, IEEE Trans. Inform. Theory, vol. 50, pp.         208-217, January 2004; or [20] A. G. Amat, G. Montrosi and S.         Benedetto, “Design and Decoding of optimal high-rate         convolutional codes”, IEEE Trans. Inform. Theory, vol. 50, pp.         267-881, May 2004.

In bit-interleaved coded modulation (BICM), coded bits are interleaved and mapped on to a transmitted symbol. Hence, there is no inner code, and the BICM mapper/modulator acts as the inner code in comparison with serially concatenated codes. Iterative decoding can be used with BICM by running iterations between the decoder and the demodulator. It is known that BICM can perform well over fading channels. Constrained interleaving can be preferably employed with BICM. When the interleaver array is formed as with serial concatenation, the coded bits can be fed along columns to the mapper. Hence, if 8-PSK is used for transmission, the interleaver array can be constructed with 3 rows (r=3) by ensuring that the coded bits of the outer code are placed in different columns. For example, if the code has minimum distance of 4, it will be guaranteed that at least 4 symbols will be different for any two transmitted sequences. In BICM applications with constrained interleaving, the constrained interleaver can be preferably constructed similar to that in SC-BC shown in FIG. 8. The optimal mapping of symbols with constrained interleaving can very well be different from that with random interleaving. Hence, it is necessary to optimize the mapping with each selected code with constrained interleaving. For example, SPC outer codes (with minimum distance 2), or Hamming codes (with minimum distance 3), or shortened Hamming codes (with minimum distance 4), or any other code can be combined with BICM using the SC-BC with constrained interleaving to improve performance.

Low-density-parity-check (LDPC) codes and related encoding and decoding thereof are known in the literature, for example, see: [21] R. M. Tanner, D. Sridhara, A. Sridharan, T. E. Fuja, D. J. Costello, “LDPC block and convolutional codes based on circulant matrices”, IEEE Trans. on Inform. Theory, vol. 50, pp. 2966-2984, December 2004; [22] M. Esmeili and M. Gholami, “Geometrically-structured maximum-girth LDPC block and convolutional codes”, IEEE Journal on Selected Areas in Communications, vol. 27, pp. 831-845, August 2009; [23] J. Kang, Q. Huang, L. Zhang, B. Zhou and S. Lin, “Quasi-cyclic LDPC codes: An algebraic construction”, IEEE Trans. on Commun., vol. 58, pp. 1383-1396, May 2010; [24] Y. Han and W. E. Ryan, “Low-floor decoders for LDPC codes”, IEEE Trans. on Commun., vol. 57, pp. 1663-1673, June 2009.; [25] M. Lentmaier, A. Sridharan, D. J. Costello, Jr. and K. Zigangiro, “Iterative decoding threshold analysis for LDPC convolutional codes,” IEEE Transactions on Inform. Theory, Vol. 56, No. 10, October 2010, pp. 5274-5289.

Constrained interleaving can also be applied to serial concatenation that involves LDPC codes. These could include a concatenation of two LDPC codes or a concatenation of an LDPC code with any other code. In the latter case, the LDPC code can be the inner or the outer code of the concatenation. For example, if a SPC outer code is used with an inner LDPC code the minimum distance of the LDPC code can be doubled with constrained interleaving and the performance of the resulting SC-BC can approach the performance lower bound given by equation (8). If two LDPC codes are concatenated with constrained interleaving, the code structure will be similar to 2-D SPC codes, and the resulting concatenation can achieve a high distance and approach the bound in equation (8). Similarly, with constrained interleaving, if an LDPC code is used as an outer code with an inner recursive convolutional code, in addition to increasing the minimum distance, the concatenation can also achieve interleaver gain. Hence, shorter and less powerful LDPC codes can be effectively used by concatenating with other codes and using constrained interleaving to generate powerful concatenated codes. In the literature LDPC convolutional codes are also known [25]. Similar to using an inner IRCC, a recursive implementation of a LDPC convolutional code can be efficiently used as an inner code along with an outer code with constrained interleaving.

It is known that LDPC codes can be decoded by considering variable nodes (also known as bit nodes) and check nodes of the code. Reviewing the literature [21-24], the variable nodes are the nodes that correspond to the coded bits while check nodes are those constructed according to the parity check equations of the code. Hence, for a (q,k) LDPC code, there are q variable nodes and a number of check nodes is equal to the number of independent parity check equations which can be derived for the given code. The Tanner graph of the LDPC code is then constructed by connecting the corresponding variable nodes to each of the check nodes according to the parity check equation of that check node.

LDPC codes are usually decoded by first assigning the soft estimates of the variable nodes from the received signals. Then the soft estimates of the check nodes are obtained using those of the variable nodes and following the connections on the Tanner graph. Then decoding is continued by running iterations between variable nodes and check nodes by exchanging extrinsic information until the stopping criterion is met or the highest allowable number of iterations is reached. In LDPC codes the stopping criterion is met when all parity check equations are satisfied. This iterative algorithm for decoding LDPC codes is referred to as the sum product algorithm (SPA) in the literature (for example, see the text book by Lin & Costello as cited in the background section herein).

Let us now focus on a serially concatenated code with a LDPC code used as a component code. For example, following the notations herein used with SC-BC, let us consider an (q,k) outer code concatenated with an (n,k) inner code and using constrained interleaving. As explained before, such an interleaver can be constructed in a q x m rectangular 2-dimensional (2-D) array. The received signal corresponding to a frame can be arranged in a n×m array with each column corresponding to a codeword of the inner code, while the coded bits of the codewords of the outer code are scattered throughout the interleaver according to the permutation policy used in the constrained interleaver. This follows the initial decoding processing as described in further detail in connection with block 515 of FIG. 5 above.

Such a code can be decoded by first loading the received sequence in an n×m array corresponding to the transmitted sequence. Then decoding can be done by directly employing the decoder shown in FIG. 5 by individually decoding the inner and the outer codes and exchanging extrinsic information through the interleaver/de-interleaver. However, noticing that the component LDPC code (when used as the inner or outer code) requires iterations within it, this direct method increases complexity. Instead, concatenated code can be more efficiently decoded by moving to the other code after a fixed number, such as one or more iterations of the LDPC code using the updated extrinsic information of the q×m array. This way the iterations of the LDPC code will be guided by the influence of the other code.

For example, consider iterative decoding of an SPC outer code with an LDPC inner code. The multiple codewords of the inner LDPC code in the q x m interleaver array are decoded using the Tanner graph of the LDPC code. After the first iteration of all inner codewords, extrinsic information is available for all array elements in the q×m array. This extrinsic information can then be used by the codewords of the outer SPC code to decode the outer code and to further update the extrinsic information of the interleaver array. Then this further updated extrinsic information can be used to run the next iteration of the LDPC decoder. Hence, in this method the outer code can be used within iterations of the LDPC code to guide the LDPC iterations. In general, when a LDPC code is used as a component code, it is possible to move to the next code after each iteration of the LDPC decoder thereby using the other code to guide the iterations of the LDPC code. This method reduces the decoding complexity compared with a direct implementation of the decoder structure of FIG. 5 where the LDPC codes are iterated until a stopping criterion is met each pass through the decoder 5. This modified decoding method can be used when at least one component code is decoded as a LDPC code.

The above decoding policy leads to the first proposed decoding algorithm, SC-LDPC Decoding Algorithm I, which can be used when at least one component code of a serial concatenation is decoded as a LDPC code. The decoding steps involved in the SC-LDPC Decoding Algorithm I can be listed as follows:

SC-LDPC Decoding Algorithm I

1. Load the received samples in an n×m array

2. Soft decode the inner code starting with the received bit metrics and using the soft decoder 515.

(a) If the inner code is a regular block code soft decode the inner code. If BCJR iterations are used, run one forward and one backward pass through the BCJR algorithm. If the regular block code is being decoded as a LDPC code, run one iteration between variable nodes and check nodes.

(b) If the inner code is the LDPC code, run one iteration (or some other fixed number of iterations) of that LDPC code decoder. In the case of an LDPC decoder, as is known in the literature (see [21]-[24]), one iteration means one update of the check nodes and coming back to variable nodes once.

Perform the appropriate operation (a) or (b) on each of the m inner codewords to obtain extrinsic information of all q×m interleaver array bits.

3. De-interleave the extrinsic information 520 to prepare for decoding of the outer code.

4. Soft decode the outer code.

(a) If the outer code is the LDPC code, run one iteration (or some other fixed number of iterations) of the soft decoder 525, using LDPC decoder. This iteration involves one update of the check nodes and coming back to variable nodes in the Tanner graph once.

(b) If the outer code is a regular block code, soft decode the outer code. If BCJR iterations are used, run one forward and one backward pass through the BCJR algorithm. If the regular block code is being decoded as a LDPC code, run one iteration between variable nodes and check nodes.

Perform the appropriate operation (a) or (b) on each of the m outer codewords to obtain updated extrinsic information of all q×m interleaver array bits.

5. Run iterations until the stopping unit 530 stops them.

6. If a next iteration through the decoder 500 is needed, constrained-interleave the extrinsic information at 535 to reorder the input to the inner decoder 515 for the next iteration.

In the above steps that involve decoding of m codewords, all m codewords can be optionally decoded in parallel to speed up decoding.

When both component codes are decoded as LDPC codes, a further modification is possible, leading to a second decoding algorithm which is referred to as the SC-LDPC Decoding Algorithm II. In such cases it is possible to consider the check nodes of the both component codes as a single set of check nodes. By doing so, both component codes can be decoded simultaneously. Then iterations can be run between variable nodes and the entire set of check nodes of all codewords of both inner and outer codes simultaneously. As a result the concatenated code is decoded similar to decoding of a single LDPC code. Therefore, this further modified decoding method will reduce the complexity to a level of decoding a single LDPC code with a number of check nodes equal to the sum of check nodes of the two component codes. As stated before, block codes can be decoded as LDPC codes. Hence, this method can be used not only when both component codes are LDPC codes but also when both component codes are decoded as LDPC codes, i.e., even when one or both component codes are block codes. For example, SPC codes can be decoded as LDPC codes, so SC-LDPC Decoding Algorithm II could be applied to a concatenation of an LDPC code with an SPC code.

In serial concatenation with constrained interleaving, the variable nodes can be arranged preferably in the n×m 2-D array as described earlier and more specifically in connection with block 515 of FIG. 5. In such a situation, as stated before, every column represents codewords of the inner codeword, while the coded bits of the m codewords of the outer code are scattered across the n×m array in accordance with the permutation policy used in the interleaver. Hence, every column represents a set of variable nodes of the inner code while the set of variable nodes of the outer code of each of the outer codewords can be identified in the n×m array in accordance with the permutation function implemented by the interleaver. The corresponding check nodes of both inner code and the outer code are then formed for each of the m codewords of the outer code and also for the codewords of the inner code. Then iterations can be run simultaneously between the entire set of n×m array of variable nodes and the entire set of check nodes both from the inner and the outer code. The iterations can be run until the stopping criterion is satisfied or until the maximum allowable number of iterations is reached. In case of LDPC decoding the stopping criterion can simply be when all parity check equations of every codeword of both inner and outer codes are satisfied, or stop when the highest allowable number of iterations is reached. The steps involved in the SC-LDPC Decoding Algorithm II can be listed as:

SC-LDPC Decoding Algorithm II

1. Load the received bit metrics on to the n×m 2-D variable node array.

2. Run iterations of between the variable nodes and check nodes for all codedwords of both inner and outer codes. Check if the stopping criterion is satisfied at the end of each iteration. Run iterations until the stopping criterion is met or the maximum number of iterations is reached.

The method and apparatus of the LDPC Decoding Algorithm II follow FIG. 5 where the soft decoders 515, 525 are implemented as LDPC soft decoders. These LDPC soft decoders are configured using variable nodes and check nodes configured as described above and as outlined in the examples and discussion below.

As an example, consider the serial concatenation of an outer SPC with an inner LDPC with constrained interleaving for high speed applications like in optical communications. The decoder can be efficiently implemented by laying out the n×m 2-D variable nodes, and placing the check nodes of both the LDPC code and the SPC code around the variable nodes. For high speed applications it is desirable to directly connect the check nodes to the corresponding variable nodes. Hence, the check nodes of the SPC codes act simply as few extra check nodes in the decoding. Specifically, since SPC codes have only one check node, the increase in the number of check nodes is only m. Hence, the increase in decoding complexity due to concatenation is minimal in this example. The connections to these SPC codes should be done based on the interleaver as the corresponding coded bits of SPC codes are scattered through the array due to interleaving. Decoding in this example is preferably performed using the SC-LDPC Decoding Algorithm II as described above.

The use of the constrained interleaver creates a natural environment to place the variable nodes and check nodes on a 2-D array. The resulting 2-D layout of the variable nodes and the check nodes in both of the above LDPC Decoding Algorithms I and II make the resulting Tanner graph of the concatenation a 2-D Tanner graph. However, depending on the application any desired number of dimensions can be used by rearranging the placement of the variable and check nodes in any desirable manner while maintaining the same connections. For example, it is also known that at high speed communications, such as in optical communications, the most efficient way to decode LDPC codes is to hard wire the appropriate connections between variable nodes and check nodes. In such situations, in order to shorten the connections from the variable nodes to check nodes, the structure of the Tanner graph can be modified in a desirable manner depending on the application using any desirable number of dimensions. It is known in the literature, once the lengths of the connections from variable nodes to check nodes increase they can cause issues which are referred to as networking issues in the literature. As noted above in the discussion of the SC-LDPC Decoding Algorithm I, different embodiments can be constructed that either move to the next decoder after a single iteration or move to the next decoder after a maximum number of allowable iterations has been performed. For example, if the LDPC decoder satisfies its own set of parity check equations after a single iteration, it can move to the next decoder. If not, it can run more iterations up to a pre-selected maximum number of iterations before moving into the next decoder.

It should be noted that even though the above LDPC Decoding Algorithms I and II are described with serially concatenated codes that use constrained interleaving, they are stand-alone inventive algorithms that can be used with serially concatenated codes that use uniform or any other type of interleaving or no interleaving at all. Likewise, these two decoding algorithms can also be used even with parallel concatenation of codes with at least one LDPC code or at least one code that employs LDPC decoding.

Like other serially concatenated codes with constrained interleaving, serial concatenation of LDPC codes with constrained interleaving with another code can significantly improve the performance of LDPC codes. This allows shorter less powerful LDPC codes to be used as component codes in the concatenation to thereby produce simpler and more powerful concatenated codes. Due to shorter LDPC component codes in the concatenation, the resulting Tanner graph can be smaller than that of an individual long LDPC code thereby reducing or eliminating the networking problems that are present with LDPC codes. Further, it is also known that iterative decoding of long LDPC codes experience undesirable error floors. The focus in the literature to combat the floor problems in LDPC codes has primarily on post-processing techniques [24]. However, as explained earlier in connection with SC-BC's, serial concatenation with constrained interleaving can eliminate these undesirable error floors. That is, the same properties of SC-BCs that solved the error floor problem apply to SC-LDPCs. Shorter LDPC codes serially concatenated with other codes using constrained interleaving can achieve high distances and generate powerful concatenated codes. These codes can be iteratively decoded efficiently, eliminate the error floor problems, and also reduce or eliminate the networking problems present with long LDPC codes.

In serial concatenation, LDPC codes can be used as outer codes where the inner code is an IRCC. Such a SC-IRCC configuration with constrained interleaving is particularly advantageous in that it can achieve a very high distance and at the same time a significant interleaver gain. Hence, this combination can be an attractive combination for many applications and embodiments as discussed hereinabove. SC-IRCCs with an LDPC outer code can be decoded by using the SC-LDPC decoding algorithm I described above. In addition, recursive implementation of LDPC convolutional codes can be used as inner codes of a SC-IRCC with constrained interleaving. In this case, LDPC Decoding Algorithm II can be used to decode the SC-IRCC that uses the LDPC as an outer code.

In the previous discussion, the SC-BC type serial concatenated codes generated in accordance with FIG. 3 use a block code for both the outer code and the inner code. The SCCC type serial concatenated codes generated in accordance with FIG. 4 use a block code for the outer code and a convolutional code for the inner code. In all of the embodiments discussed thus far, the constrained interleaver's permutation function implements a constraint in order to enforce d_(i)<d_(sc)≦d₀d_(i). The distances d_(sc), d₀ and d_(i) can be representative of Hamming distances. In particular, for the specific examples provided above, the constrained interleaver's permutation function implements a constraint that enforces d_(sc)=d₀d_(i).

When the inner code is a block code or a non-recursive convolutional code, the highest achievable MHD of the concatenation is d₀d_(i) which is achieved by the examples provided above. However, as discussed in F. Danesgaran, M. Laddomada and M. Mindin, “Interleaver design for serially concatenated convolutional codes: Theory and application”, IEEE Trans., IT-50, No. 6, pp. 1177-1188, June 2004, (“the Daneshgaran reference”) when the inner code is a recursive convolutional code, the MHD of the concatenation can in fact be increased beyond d₀d_(i). Herein, a constrained interleaver that is designed as hereinabove to enforce d_(sc)=d₀d_(i) is referred to as a “constrained interleaver type 1” or “CI-1.” For the case of SCCC's (the inner code is a recursive convolutional code) additional constraints can be added to enforce the MHD of the concatenation to be increased beyond d_(sc)=d₀d_(i) so that d_(sc)>d₀d_(i). Herein, a constrained interleaver that is designed to enforce d_(sc)>d₀d_(i) is referred to as a “constrained interleaver type 2” or “CI-2.” A constrained interleaver that trades off distance for interleaver gain to achieve d_(i)<d_(sc)<d₀d_(i) is referred to as a “constrained interleaver type 0” or “CI-0.”

The Daneshgaran reference designs interleavers for serial concatenated codes where both the outer code and the inner code are recursive convolutional codes. The interleavers of the Daneshgaran reference are structured to iteratively expand themselves by focusing on minimizing a cost function based on the error contributions by different patterns. Note that a convolutionally encoded sequence, if viewed as a codeword of a block code, is a single codeword. This is different from constrained interleaving where the outer code is a block code, and as such, there are multiple codewords present in the constrained interleaver as opposed to a single large codeword as in the Daneshgaran reference. The iterative interleaver design method of the Daneshgaran reference starts with a set of initial problematic error events that can reduce the minimum distance. Once the interleaver is expanded, the algorithm aims to re-position the bits involved in these error events with the aim of reducing the error contributions made by them. To iteratively move to a larger interleaver size, a set of new possible error patterns are derived from the previous set of error patterns (before the expansion of the interleaver size) and the new positions are determined based on the cost function (which is the total error probability contributions).

This interleaver design method of the Daneshgaran reference is a refinement of S-interleavers and is based on a pre-selected SNR. This interleaver design method of the Daneshgaran reference would work only when the entire interleaver is filled by one codeword of a code. Hence, if applied to a serially concatenated code whose outer code is a block code, the interleaver design method of the Daneshgaran reference would fail as the error events with lower distances, when the interleaver length is increased, cannot be determined from those of lower interleaver lengths. If the algorithm is attempted to be modified to even include all the new error events introduced when new codewords are added one at a time, the additional important error events would increase very rapidly and such an interleaver design technique is not practical.

Constrained interleaving, on the other hand, is used to construct serially concatenated codes. The serially concatenated codes constructed with constrained interleaving use an outer code that is a block code (or a finite length convolutional code). The inner code used in the serially concatenated code may be selected to be either a recursive convolutional code or a block code (or a non recursive convolutional code). Because the outer code used with constrained interleaving is a block code, there will be multiple codewords present in the constrained interleaver. This is in contrast to the interleavers of the Daneshgaran reference, where the outer code is a recursive convolutional code, so that the bits inside the interleaver correspond to a single long codeword.

The interleavers of the Daneshgaran reference are not constrained interleavers as contemplated by the present invention because they do not implement interleaver constraints as taught by the present invention. Constrained interleaving can be summarized in that it: 1) uses an outer code that is a block code or a non-recursive convolutional code, and as such, there are multiple codewords present in the constrained interleaver, 2) selects a desired MHD, 3) selects an interleaver size and a set of predefined interleaver constraints to prevent undesired error events so as to achieve the desired MHD, and 4) performs uniform interleaving among all or at least a subset of the allowable (non-constrained) positions, to thereby maximize or otherwise improve the interleaver gain subject to the constraints imposed to maintain the desired MHD.

Constrained interleaving teaches the way to design CI-1 and CI-2 interleavers in serial concatenation when the outer code is a (n,k) block code with MHD d₀ and the inner code is a recursive convolutional code with free distance d_(i) to achieve a MHD of the concatenation d_(sc)≧d₀ d_(i). Also, constrained interleaving teaches the way to design interleavers in serial concatenation when the outer code is a (n,k) block code with MHD d₀ and the inner code is also a block code or a non-recursive a convolutional code with free distance d_(i) to achieve a MHD of the concatenation d_(sc)=d₀d_(i). In both cases, constrained interleavers could optionally also be designed to achieve d_(sc)≦d₀d_(i). This option can be useful if the effects of the error coefficient are contributing more to the desired BER than the MHD.

As is discussed in more detail below, CI-2's can be designed for SCCCs to enforce d_(sc)>d₀d_(i) by increasing the number of rows over what is required for a CI-1 design, and by further imposing inter-row constraints among rows. In order to simplify the description, we develop specific embodiments by way of example. For example, a preferred embodiment uses an inner rate-1 code, however, the same concept can be applied to design constrained interleavers for other inner codes as well.

For example, it can be noted that a CI-1 of an SPC outer code that has d₀=2 and a rate-1 inner code that has d_(i)=1 achieves a MHD of the concatenation d_(sc)=d₀d_(i)=2. Further, when the inner code is a rate-1 inner code that has unit memory, the required number of rows of CI-1 interleaver is r=2. The MHD of this concatenation can be increased to 4 by increasing the number of rows to 4 and ensuring that coded bits of any two codewords on adjacent rows share no more than one common column. These inter-row constraints can be extended to include rows beyond the immediately previous row. In general, the assignment of coded bits on any i^(th) row can be made dependent on up to l_(max) number of previous rows. In this structure, due to the cyclic nature of feeding bits into the inner code by going back to the first row after the r^(th) row in the next column, the placement of bits on any (r−i)^(th) row (i<l_(max)) depends not only on the l_(max) previous rows but also the first (l_(max)−i) rows. In general, any specific inter-row constraint on the i^(th) row for 1<i<(r−l_(max)) can be expressed as: coded bits of any codeword on the i^(th) row cannot share no more than k(l) common columns with coded bits of any codeword placed on the (i−l)^(th) row, where l=1, 2, . . . , l_(max). The value of l_(max) and the set of values k(l)<d₀, l=1, 2, . . . , l_(max), define all inter-row constraints of the CI-2. As stated before the set of inter-row constraints is often imposed in a cyclic manner meaning that the inter-row constraints on the k^(th) column of the i^(th) row, for (r−l_(max))<i<r, comes not only from the k^(th) column of (i−l)^(th) row but also from the (k+1)th column on the (l_(max)−r+i)^(th) row, for 1≦l<l_(max). Along with the values of r and ρ that were used also in CI-1, the set of inter-row parameters, l_(max) and k(1),k(2), . . . , k(l_(max)) define a CI-2. The target MHD of the concatenation, d_(t), can be made equal to 2d₀d_(i) with l_(max)=1. However, when d_(t)>2d₀d_(i), in addition to the inter-row constraints, a set of intra-row constraints can be used to avoid placement of non-zero coded bits of one or more valid codewords of the outer code with total weight less than [d_(t)/l] on the i^(th) row in the same columns as with those of one or more valid codewords on the (i−l)^(th) row with the same total weight, where └x┘ denotes the floor function of x. As with CI-1, CI-2 allows any coded bit to be placed anywhere in the interleaver array, however, due to the inter-row constraints, the flexibility is limited as compared to a CI-1 design.

A CI-2 can be systematically constructed by placing coded bits of ρ codewords (n₁ρ bits) on a row, one row at a time. The first row can be filled in any random order by the coded bits of any randomly selected set of ρ codewords from the entire set of rρ codewords. However, all other remaining rows, 2 through r, need to be filled according to the inter-row constraints. In order to enhance the chances of finding a CI-2 that satisfies all inter-row constraints, it is desirable to fill coded bit positions of all ρ codewords one by one up to the n₁ ^(th) position. When filling any i^(th) row, the first k_(min)=min_(l) k(l) coded bits of all codewords can be randomly placed anywhere on the row as they alone cannot violate any inter-row constraint. Coded bits of any other bit position of all codewords can be placed one bit at a time starting from the first codeword up to the ρ^(th) codeword. Any such coded bit can be placed by eliminating columns according to the inter-row constraints for each of those bit positions one at a time. For example, if l_(max)=1 and k(1)=1, the placement of the second coded bit of any codeword on the second row can be selected by disregarding the n₁ columns used by the codeword on first row that share a column with the already placed first coded bit of that codeword on the second row, and then randomly selecting a position among the remaining n₁(ρ−1) positions. In failing to fill according to inter-row constraints, any row can be tried multiple times until a valid placement that agrees with all inter-row constraints is found. The last row, which requires consideration of inter-row constraints from rows (r−l_(max)) through (r−1) and rows 1 through l_(max), determines the minimum requirement on ρ. When d_(t)=2d₀d_(i) (for which l_(max)=1), realizing that some of the columns removed due to inter-row constraints from rows (L−l_(max)) through (L−1) and rows 1 through l_(max) can be the same, the last (n₁ ^(th) bit) of a codeword on the last L^(th) row requires elimination of at least

$\left\lfloor \frac{n_{1}}{k(1)} \right\rfloor$ columns. Considering filling up the last row, a CI-2 that satisfies all inter-row constraints when d_(t)=2d₀d_(i) can be successfully found with a value of ρ≧n₁, and a valid CI-2 can be numerically found with a value of ρ close to n₁. However, when d_(t)>2d₀d_(i), due to the presence of additional intra-row constraints, the required value of ρ increases beyond that of d_(t)=2d₀d_(i). Further, as with CI-1, the interleaver gain of any CI-2 can be increased by increasing the value of ρ. Since the value of ρ grows with n₁, in order to limit the size of the interleaver N=Lρn₁, the CI-2 technique is attractive for small to medium size outer codes whereas CI-1 can be used with any size of an outer code with integer ρ.

The MHD of CI-2, d_(min)(CI-2), can be bounded by the parameters of the interleaver. By considering the worst case distance generated by a single non-zero codeword of the outer code with weight d₀, and two codewords of the outer code each with weight d₀, d_(min)(CI-2) can be bounded respectively as

$\begin{matrix} {{d_{t} \leq {r\left\lfloor \frac{d_{o} + 1}{2} \right\rfloor}},{d_{t} \leq {d_{o}\left( {l_{\max} + 1} \right)}}} & (64) \end{matrix}$

Similarly, d_(t) can be bounded according to the inter-row constraints by considering two codewords of the outer code each with weight d₀ on rows i and (i−l) as d _(t) ≦lk(l)+[d ₀ −k(l)](r−l), l=1, 2, . . . , l _(max)  (65)

Hence, for a given outer code with a known d₀, the parameters r, ρ, l_(max) and k(1), k(2), . . . , k(l_(max)) can be selected to maintain a desired MHD for the concatenation according to (64) and (65).

As explained above, a CI2 is described by its parameters, r, ρ, l_(max), and the set of values k(l) for l=1, 2, . . . , l_(max). These parameters are usually selected according to Eqns. (64), (65) and (66) to limit the interleaver size and to achieve a desired target distance.

By way of example, the CI-2 interleaver's permutation function can be more specifically designed by starting with a r×n₁ρ matrix of bit positions and taking the actions summarized below:

3. Randomize a length-rρ Input_Block of n_(l)-bit codewords (CW's).

Rand_Input_Block=Rand_CW (Input_Block),

where Rand_Input_Block denotes a uniformly interleaved set of n_(l)-bit codewords of the outer code after randomizing, and Rand_CW denotes the uniform interleaving operation applied to randomize n_(l)-bit codewords as opposed to bits. This action arranges all rρ codewords in a random order.

4. Select the first ρ codewords from the randomized set of rρ codewords, place these first ρ codewords in the first row and randomize its contents according to Rand_Row₁=RandRow₁(Row₁)

where, Rand_Row₁ denotes the contents of the first row after randomizing, and RandRow₁ denotes the uniform interleaving operation used to randomize the contents of the first row.

At this point, we move to the next (in this case second) row. In general, let us consider the placement of coded bits on the i^(th) row.

5. Select the next set of p codewords from the list created in step 1. Select the first k_(min)=min_(l=1 . . . l max){k(l)} bit positions of all selected p codewords. Randomly place these ρk_(min) bits on the ith row. At this point (n₁ρ−n₁k_(min)) columns of the ith row are available for further bit mappings. Note that this guarantees that no coded bits mapped thus far to the i^(th) row will share more than k_(min) columns on row (i−l) for l=1 . . . l_(max).

The following actions 4-9 determine valid bit mappings (permutations) for coded bits on the i^(th) row, advancing one bit position at a time for all ρ codewords on the i^(th) row, starting from the (k_(min)+1)th position up to the n_(l) ^(th) position. Valid bit mappings are bit-wise pseudo random permutation rules (bit mappings applied to coded bits on a row) that ensure the inter-row constraints are satisfied.

6. For the placement of the k^(th) bit of the j^(th) codeword on row i there will be [ρ(n₁−k+1)−(j−1)] columns still available for an unconstrained pseudo random mapping of this bit. However, in order to ensure the inter-row constraints are met, some of these columns may need to be avoided. Hence we identify a set, C_(IR) that identifies the columns that need to be avoided to meet the inter-row (IR) constraints when mapping the k^(th) bit of the j^(th) codeword on row i. To identify the set, C_(IR), first define the set of column indicies C=(c₁, c₂, . . . , c_(k-1)) corresponding to the columns occupied by the coded bits 1 through (k−1) of the jth codeword already placed on the ith row. For l=1 . . . l_(max), look at row (i−1) and determine if one or more of the codewords already placed on row (i−1) have k(l) coded bits mapped to the columns whose indicies are specified by C. Identify all such codewords on row (i−l) that have k(l) coded bits already mapped to the columns whose indicies are specified by C, and form a set C_(s)(l) of the column indicies of the remaining (n−k(l)) coded bits from all such codewords. Note that if all codewords on row (i−l) have less than k(l) coded bits mapped to the columns whose indicies are specified by C, the set C_(S)(l) will be empty. The determination of C_(S)(l) is also referred to as checking on the (i−l)th row herein.

Once all C_(S)(l) sets are found for l=1, 2, . . . , l_(max), let C_(IR)=∪_(l=1 . . . l max)C_(S)(l). Note that C_(IR) contains all column indices found from each row (i−l)th row for l=1, 2, . . . , l_(max).

7. Randomly map the kth coded bit of the jth codeword of the ith row to a column index from the available [ρ(n₁·−k+1)−(j−1)] columns while avoiding any column index contained in the set C_(IR).

8. Move to the next codeword (j+1) at the same bit position k and repeat steps 4 and 5 until the kth bit position of all ρ codewords are placed on that ith row.

9. Once the kth bit position is complete, move to the next (k+1)th bit position and repeat steps 3 through 6 until the n₁ ^(th) bit position on the ith row has been placed.

10. Move to the (i+1)th row.

11. Repeat steps 3 though 8 until to the (r−l_(max))th row is reached.

For each of the rows (r−l_(max))<i≦r, step 4 has be expanded to also include the effect of the top (l_(max)+i−r) rows. First, by following step 4 exactly the way it has been described, find the sets C_(S)(l) by checking rows (i−l), for l=1, 2, . . . , l_(max). In addition, for all rows (r−l_(max))<i≦r, it is also necessary to check the first (l+i−r) rows at the top of the interleaver. Specifically, similar to checking with the row (i−l), the row (l+i−r) has to be checked against the same value k(l). However, when checking with row (l+i−r), for each l=(r−i+1), (r−i+2), . . . , l_(max), all row indices in C have to be increased by one to form a modified C, C_(mod). This is because on the top rows, the column where the k^(th) bit will be placed will interfere with the bit positions one column ahead. If any of the indices of C_(mod) are above n₁ρ, these elements can be dropped from C_(mod). Using C_(mod) and k(l), row (l+i−r) is checked by following step 4. In order to differentiate from the set C_(S)(l) that has already been found from row (i−l), the set C_(S)(l) found in step 4 for row (l+i−r) is denoted by C_(Q)(l). All column indices of C_(Q)(l) are reduced by one to form the new set C_(F)(l). By following step 4 with the C_(mod), all C_(F)(l), l=(r−i+1), (r−i+2), . . . , l_(max) are found. All column indices of C_(F)(l) are first reduced by one. If any index in any modified C_(F)(l) is zero, drop that index from C_(F)(l). The set C_(R) is then found from the union of all sets C_(S)(l) l=1, 2, . . . , l_(max) and C_(F)(l) for l=(r−i+1), (r−i+2), . . . , l_(max). Follow steps 5 through 8 down to the last (rth) row. On each of the ith row below the (r−l_(max))th row, in addition to the sets C_(S)(l) for l=1, 2, . . . , l_(max), the sets C_(F)(l) l=(r−i+1), (r−i+2), . . . , l_(max) have to be calculated, and they all then contribute to the set C_(R) as explained above.

The above steps 1 through 9 can be used directly to design a CI-2 interleaver to achieve a target distance of d_(t)=2d₀d_(i). However, if the target distance is d_(t)>2d₀d_(i), steps 4 and 5 should be modified to include the intra-row constraints too. As stated before, the intra row constraints do not allow placement of one or more valid codewords with a total weight less than └d_(t)/l┘ to be placed on the ith row that share the non-zero coded bits in the same column indices as with another combination of one or more codewords with the same weight └d_(t)/l┘ on the (i−l)th row. Step 4 should be modified to expand the set C_(IR) that would include all columns that that the kth bit of the jth codeword cannot be placed on the ith row. When d_(t)>2d₀d_(i), a CI-2 interleaver can also be designed by following the above steps 1 through 9 without any modifications (i.e., disregarding the intra-row constraints), and then using the following alternate method to modify the interleaver, if necessary, to satisfy the intra-row constraints.

Once the above design process has been completed, a set of constrained pseudo-random permutations will have been defined. The first permutation is applied to a block of codewords and is defined as, Rand_Input_Block=Rand_CW (Input_Block). At this point the bits of the permuted code words can be loaded in row-major order into a r×n_(l)ρ matrix of bits. Next the rows are permuted in accordance with Rand_Row₁=RandRow₁(Row₁), where this permutation is an unconstrained row permutation as defined above, and Rand_Row₂=RandRow_(IR-2)(Row₂) . . . Rand_Row_(r)=RandRow_(IR-r)(Row_(r)) are the resulting permutations for each of the remaining rows that map coded bits to additionally meet the inter-row constraints as defined by the bit mappings above. Bits are read out of the interleaver matrix in column major order. That is, at run time, the constrained interleaver just maps the bits according to pre-designed pseudo-random permutation rules that meet the constraints of the constrained interleaver as determined by the design procedure above. Hence this constrained interleaver is no more costly to implement than the previously described constrained interleavers. The only difference is the bit-wise pseudo-random permutation rules are preselected to ensure that the inter-row constraints are met. For example, constrained interleavers can be designed using the CI-1 or CI-2 design techniques as discussed above, or a CI-2A design technique as discussed below, or a similar design technique (e.g., a “CI-X”) that insures a selected constraints of interleaver constraints are met.

To better understand the operation of the CI-2 in operation at run time, consider FIG. 13. The CI-2 operates similarly to the CI-1 type interleaver 1300. The only difference is that the row permutations of block 1315 are implemented to ensure the inter-row constrains are met. Similarly, for the CI-2A as described below, the only change is that the row pseudo-random permutation functions of 1315 are selected to meet the type 3 constrains. More generally, the CI-1 1300 embodiment can be modified to add additional constraints which further limit the pseudo-random row permutations of the block 1315.

The CI-2 can be alternatively implemented in various ways that assure the inter-row constraints are met. For example, an alternative interleaver design embodiment starts by first selecting each row randomly and then making adjustments if necessary to satisfy the inter-row constraints. In this alternative design approach, rows can again be selected one at a time down to the last (rth) row. We can again use steps 1 and 2 above, but the other steps would be modified. In order describe these modifications, let us consider the selection of any ith row (1<i<=(r−l_(max))) as follows:

3. Randomly assign positions for n₁ bits of all ρ codewords on the ith row. In other words, random interleave all n₁ρ bits on the ith row.

At this point, there can be violations according to inter-row constraints. Hence, we need to first check to see if there are violations; and if there are, identify them, and correct them, and if not accept that assignment and move to the next row. For checking, identifying and correcting, for example, we can use the following steps:

4. Prepare a ρ by n₁ array A, that lists the codeword numbers (1 through ρ) of the codewords on rows (i−1), (i−2), . . . , up to the (i−l_(max))th or the first row whichever comes first that share columns with each of the coded bits of codeword on the ith row. Each of the (j,k) entry on the array A (j=1, 2, . . . , ρ, k=1, 2, . . . , n₁) can be expressed as a vector p_(j,k)=(p₁, p₂, . . . p_(max)), where p_(max)=Min{(i−1), l_(max)}. Note that each p_(l) of the vector p_(j,k) represents which codeword (1 through ρ) on the interleaver row (i−l) that shares a row with the kth coded bit of the jth codeword placed on the ith row of the interleaver. The array A carries enough information to check for any violations to inter-row constraints and to identify their locations in case there are violations. Specifically, on any jth row of the array A, if there are more than k(l) common entries at the positions p_(j,k) along all k values for a given codeword j, the inter-row constraint imposed k(l) has been violated. By examining the array A, identify the set of all codewords on the ith row of the interleaver that violates the inter-row constraints. If there are no violations accept the current selection of the ith row of the interleaver and move to the next row. However, if there are violations move to step 5 to correct them.

5. In order to make changes on a row that has failed to pass the checks on inter-row constraints, we add another entry besides the vectors p_(j,k) in the array A to identify the column occupied by the kth coded bit of the jth codeword on the ith row of the interleaver. Hence, the array A has ρ rows and n₁ columns, and every (j, k)th entry that corresponds to the kth coded bit of the jth codeword has a vector p_(j,k) and a position index pos_(j,k) that indicates its column number. For all problematic codewords identified in step 4, it is necessary to swap entries on the array A to eliminate the violations. These swaps can be identified by considering one problematic codeword at a time and finding a good candidate entry on A to swap it with to resolve the violation without however initiating new violations. The candidates to, swap can be searched by tracing through the array A (say from top to bottom) and finding the first acceptable position to swap for each problematic entry. The effect of each swap can be checked as explained in step 4. Record all the swaps that have been made in the original array A to arrive at the new array A that has eliminated violations to all inter-row constraints. Since every entry carries its pos_(j,k) values, swapping two entries (say the (j₁, k₁) entry with the (j₂, k₂) entry) on A is identical to swapping the corresponding two positions indicated by the two pos_(j,k) values (swapping pos_(j1,k1) with pos_(j2,k2)) on the ith row of the interleaver. Hence, by following all swaps that have been made in the array A in order to eliminate all violations, swap the bit positions on the current ith row of the interleaver corresponding to all swaps made on the array A using their respective pos_(j,k) values.

6. Follow steps 3 through 5 down to the (r−l_(max))th row.

7. For all the remaining l_(max) lower rows, as in the previous method, it is necessary to consider the effect of the top rows too in checking the inter-row constraints. Specifically, when working on the ith row of the interleaver, for (r−l_(max))≦i≦r, it is necessary to extend the vector p_(j,k) in the array A to include the codewords numbers on rows 1 through (l+i−r) that share columns with those on the ith row of the interleaver. As in the previous method, it is understood that sharing with a column c on the ith row mean the column (c+1) on any of (l+i−r) top rows. With this modification and expanding p_(j,k) to include the top rows, steps 4 through 6 can be followed to complete the filling the lower (r−l_(max)) rows to complete the interleaver.

Again, the above design procedure is used to find a set of pseudo-random bit-permutation functions that can be applied to each respective row at run time. For example, at run time, for each block of codewords, as described above, the codewords are first permuted, then the bits are read into the interleaver array in row major order, the row permutation functions are applied, and bits are read out of the interleaver array in column major order. The interleaver permutation function can be implemented in various ways, but this way is considered to be most desirable at this time. Alternative implementations could use arrays of pointers or similar data structures to implement the permutation function similarly.

The CI-2 embodiments above have been described by considering additional constraints in the form that a codeword on row i cannot share more than k(l) number of coded bits with a codeword on row (i−l). Additional constraints of that type are suitable for SPC type outer codes. However, for other outer codes with higher minimum distances, the additional constraints can be custom tailored depending on the selected pair of outer and inner codes. That is, the inter-row constraints can be replaced more generally by “additional constraints” that are selected to maintain a minimum distance or to otherwise jointly account for minimum distance and the effect or error coefficient. It should be noted that the constrained interleavers of the present invention make use of interleaver constraints that can be implemented as described herein because the outer code is a block code or some other type of non-recursive code. The constraints used to design constrained interleavers in accordance with the present invention are possible since the outer code is a block code or some other type of non-recursive code.

The present invention further contemplates that constrained interleaving can be employed within each stage of a multiple concatenation. One of the techniques, CI-1 or CI-2, can be employed in each stage if the component code that immediately follows the interleaver in that stage is a recursive code. A preferable choice would be to employ CI-2 in one or more early stages with the focus on increasing the minimum distance. Next a CI-1 would be used in the later one or more stages with the focus on increasing the interleaver gain. For example, in a preferred embodiment involves a double concatenation of an outer block code with two recursive codes and two constrained interleavers. A CI-2 is preferably placed between the first and the second component codes, and a CI-1 is preferably placed between the second and the third component codes. That is, this preferred embodiment uses an outer block code with two recursive codes and a CI-2 in the first interleaver and a CI-1 in the second interleaver. Further, in order to maximize the interleaver gain, the second interleaver can be preferably filled by independently generated first concatenations that preferably employ independent CI-2 interleavers.

FIG. 18 shows and exemplary embodiment of a double SCCC that has two inner codes, an IRCC-1 1815 and an IRCC-2 1825. In order to simply the explanation of the embodiment, we consider two rate-1 codes for IRCC-1 and IRCC-2 in FIG. 18. This embodiment consists of a double concatenation, where the first concatenation is formed by the (n,k) block code as the outer code and the first rate-1 code as the inner code, while the second concatenation is formed by the first concatenation as the outer code and the second rate-1 code as the inner code. As illustrated in FIG. 18, all codewords of the first concatenations 1805, 1810, 1815 . . . 1805, 1810A, 1815 are generated in parallel. In a preferred embodiment, the constrained interleavers 1810, . . . 1810A are implemented as with independently designed CI-2 constrained interleavers that are each independently designedto maintain a common desired target distance. The constrained interleavers 1810, . . . 1810A will have different permutation functions because the pseudo-random aspects of these interleavers will be different each time the CI-2 design algorithm is run. To form a second concatenation r₂ρ₂ codewords of the first concatenation are placed in the second interleaver 1820. The second interleaver 1820 is preferably designed according to CI-1. In the exemplary embodiment of FIG. 18, the second interleaver 1820 has r₂ rows, and ρ₂ codewords of the first concatenation are placed along each row of the second interleaver 1820. The second interleaver 1820 is implemented to have r₂ rows in it. By placing ρ₂ such codewords on each row of the second interleaver 1820 the dimensions of the interleaver 1820 become r₂ by r₁ρ₁ρ₂n. The exemplary double concatenated code as generated by the encoder of FIG. 18 will be a (r₁r₂ρ₁ρ₂n, r₁r₂ρ₁ρ₂k) code with rate k/n.

In operation, a single frame includes r₁r₂ρ₁ρ₂k message bits. These bits are first grouped into r₂ρ₂ groups, each with r₁ρ₁k bits. As illustrated in FIG. 18, each group is then processed in parallel by feeding its message bits into the outer (n, k) block code k bits at a time, to generate n bit codewords. Each parallel branch processes r₁ρ₁ such codewords in its interleaver according to CI-2 interleaving. Each parallel group then feeds the contents of its CI-2 interleaver into its rate-1 code to generate its codeword of the first concatenation. In total, each of the codewords of the first concatenation is r₁ρ₁n bits long. These codewords are then placed in the second interleaver according to CI-1 rules. The final coded output bit stream of the double concatenation (which is also the output of the second concatenation or the second rate-1 code) is r₁r₂ρ₁ρ₂n bits long.

Other embodiments can be found by changing the component codes and the type of interleaving. The parameters of the interleavers, the codeword lengths of the first concatenation and the overall double concatenation change according to the selected changes. These changes can include one, several or all of the following:

(a) different combinations of CI-1, CI-2 and CI-2A techniques in the two interleavers

(b) Different IRCC codes other than the rate-1 code

(c) Use of a block code as the first IRCC and change the interleavers in the parallel branches to follow rules of CI when the inner code is a block code according to the three steps described in connection with the constrained interleaver 310 when used with the of the (q, k) outer code and the (n, q) inner code to form the (mn, mk) serially concatenated block code in the SC-BC embodiment of FIG. 3.

(d) Instead of using r₂ρ₂ parallel groups, each with an independent interleaver of the first concatenation, one common first concatenation can be used. In this case the same identical first concatenation that generates r₁ρ₁ long codewords each time will be used r₂ρ₂ times before completing the second interleaver.

A variation of CI, called CI-2A (constrained interleaver type 2A), is also discussed here with the intention of increasing the MHD of a concatenation of an outer block code and an inner recursive code. As CI-2, CI-2A is explained with a rate-1 inner recursive code, however, the inventive design concepts provided herein can be readily applied for other inner recursive codes as well. We note that the CI-2A is a special case of a CI-2 but one that uses a different type of interleaver constraint to achieve d_(sc)>d₀d_(i). The CI-2A constraints are even more restrictive than the CI-2 constraints discussed above and can thus target higher values for d_(sc). In general, other specific constraints could also be developed and still fall within the class of CI-2 as long as the constraints are selected to enforce d_(sc)>d₀d_(i). It is preferred that once the constraints are met, the interleaver's permutation function is selected to be pseudo-random among the non constrained positions in order to achieve a high interleaver gain at the same time as meeting the distance constraints.

In CI-2A, the coded bits of all codewords in a frame are fed into the inner code one coded bit position at a time. Each bit position i of all codewords in the frame are arranged in a row column array with r_(i) rows and fed along columns into the inner code. Hence, every ith coded bit position is essentially row/column interleaved with r_(i) rows. The set of values r_(i) for i=1, 2, . . . , n_(l) can be chosen in the ascending order based on the desired MHD and the allowed interleaver size. If necessary, r_(I) can be chosen to be one. For an outer code with MHD d₀ and a rate-1 inner code, the achievable MHD of the concatenation can be bounded by

$\begin{matrix} {d_{\min} \geq {\sum\limits_{i = 1}^{d_{0}}\; r_{i}}} & (67) \end{matrix}$

Depending the MHD of the outer code, the MHD of the concatenation can possibly be adjusted to be the sum of d₀ number of lengths r_(i), different 1 through d₀, to further increase MHD of the concatenation.

Further, since all row/column interleavers need to have the same integer number of columns on each row, the number of codewords in a frame, ρ_(t), has to be at least ρ_(t)=LCM{r₁, r₂, . . . , r_(n)}, where LCM stands for the least common multiple. Note that, in terms of interleaver size, ρ_(t) is equivalent to rρ in CI-1 and CI-2. Depending on the outer code, in order to maintain the MHD given by (67), it may be required to add a certain number of all zero codewords (say x) at the end of ρ_(t) codewords in a frame that guarantees transmission of x zeroes after completing every coded bit position before moving to the next position. Despite a slight reduction in the rate, this may be required to avoid special merging events with lower distances depending on the outer code.

A CI-2A for an (n,k) outer code and a selected set of row lengths {r₁, r₂, . . . , r_(n)} is constructed to transmit coded bit of all ρ_(t) codewords in a frame one bit position at a time. A CI-2A can be designed according to the following steps:

(a) select an appropriate value of ρ_(t) according to ρ_(t)=LCM{r₁, r₂, . . . , r_(n)}

(b) group all ρ_(t) bits of each bit position from 1 through n

(c) use a row/column interleaver with r, rows for the ith bit position, i=1, 2, . . . , n

(d) If necessary, add a desired number of x all zero codewords to the set of ρ_(t) codewords.

Variations of CI-2A can include the same value of r_(i) multiple number of times in the different row/column interleavers. Further, instead of feeding contents of the interleavers along columns into the inner code in all inetrleavers, contents can be fed into the inner code according to different patterns in different interleavers especially when the same value of r_(i) is repeated. These directions can include North East, North West, South East, South West directions or according to any other pattern.

It is noted that the decoder structures such as 500 and 600 developed for codes concatenated using a CI-1 can also be applied to codes concatenated using CI-2 (and CI-2A). Of course, the decoder will need to use the new mapping policy determined by the CI-2 (or CI-2A) interleaver, but other than that the decoding is done by iterative decoding as previously discussed. When exchanging extrinsic information the decoders 500 and 600 should use the same permutation function that is implemented by constrained interleaver in use in the encoder. The permutation function used by the constrained interleaver can be designed using any desired interleaving policy (e.g., CI-0, CI-1, CI-2 or CI-2A, etc). When exchanging extrinsic information, the decoder needs to use the same permutation function as used in the encoder.

As taught in the Narayanan reference, concatenated codes can be alternatively decoded as low density parity check (LDPC) codes using the sum product algorithm (SPA). It is known that LDPC codes can be decoded by exchanging information between data points (for received bits) and check points. The SPA algorithm can be employed to decode concatenated codes by considering multiple sets of data points corresponding to the contents of each interleaver, and the final coded bits of the concatenation. The check points can then be formed by considering appropriate data points from one or more data point sets. The data point and the check point structure (the Tanner graph [see the Shu Lin reference]) can be alternatively used even with the BCJR decoding of component codes to obtain stopping criterion. The iterations can be run, one at a time or a pre-selected number of them at a time, until all check equations are satisfied.

The receiver can be alternatively constructed to reduce the complexity (however, at the expense of performance) by hard decoding each component code of the concatenation instead of soft decoding. When hard decoding is used, most current hard decisions available on the contents of the interleaver can be directly used in the decoding of the next component code. The iterations should be run until the contents of the interleaver remain unchanged pointing that the iterations have reached a valid solution. If the iterations fail to reach such a valid solution, the received signal can be modified (perturbed) until the solution of the first component code in the first iteration is different from the previous set of iterations, thereby creating a new starting point for a new set of iterations. The received signal can be modified by observing the bit positions that alternate during iterations that causes an invalid solution and then use those bit positions to decide on the perturbation in the received signal that need to introduced so that it causes a minimum change in the Hamming distance, or Euclidean distance, or any such measure from the actual received signal.

Although the present invention has been described with reference to specific embodiments, other embodiments may occur to those skilled in the art without deviating from the intended scope. Figures showing block diagrams also identify corresponding methods as well as apparatus. All “transmitted signals” shown in the Figures can be applied to various types of systems, such as cable modem channels, digital subscriber line (DSL) channels, individual orthogonal frequency division multiplexed (OFDM) sub-channels, and the like. Systems can be configured whereby a transmitter sends information to a receiver, for example on a wireless OFDM channel used in WiFi and WiMAX systems. In general, more than two component codes can be concatenated together, and embodiments can be created that mix parallel and serial concatenation to form mixed parallel/serial concatenated codes. In such cases the constrained interleaving can be performed on any component-encoded or concatenated encoded bit stream to be interleaved within the mixed encoder structure to satisfy a constraint that is designed to jointly optimize or otherwise improve bit error rate performance by jointly increasing a measure of minimum distance and reducing the effect of one or more dominant error coefficients of the mixed encoded bit stream. The concepts presented herein can be extrapolated to these higher order cases by induction. The present invention can generate coded schemes that eliminate the undesirable error floor effects present in known serial and parallel concatenated codes. This attractive property makes serial concatenated codes with constrained interleaving a potential coding scheme for low error rate applications such as in optical communications and in magnetic recording. Hence it is noted that all such embodiments and variations are contemplated by the present invention. 

What we claim is:
 1. A communications apparatus comprising: a constrained interleaver configured to receive a sequence of N=r₂m input bits that corresponds to r₂ρ₂ number of multibit codewords that were generated from r₂ρ₂ number of independent encoders, and the constrained interleaver is further configured to rearrange an ordering of the sequence of N=r₂m input bits in accordance with a permutation function in order to produce a permuted sequence of output bits, wherein N, r₂, ρ₂, and m are positive integers and the rearrangement of the ordering in accordance with the permutation function is equivalent to performing the following operations: loading a r₂×m array of bits, wherein the r₂×m array has r₂ rows and m columns, and the bits are serially loaded into the r₂×m array of bits along rows, applying an i^(th) pseudo-random row permutation function to each row i, for i=1,2, . . . r2, wherein the i^(th) pseudo-random row permutation function pseudo-randomly rearranges an ordering of the bits in the i^(th) row, applying a j^(th) pseudo-random column permutation function to each r₂-bit column j, for j=1,2, . . . m, wherein the j^(th) pseudo-random column permutation function pseudo-randomly rearranges an ordering of the bits in the j^(th) r2-bit column, and reading bits out of the r₂×m array along r₂-bit columns to form the permuted sequence of output bits; wherein: each of the r₂ρ₂ number of independent encoders encodes in accordance with a respective serially concatenated code, the first respective component encoder is an outer encoder configured to encode in accordance with an outer code that has a Hamming distance of d₀, the second respective component encoder is an inner coder configured to encode in accordance with an inner code that has a minimum Hamming distance of d_(i), coded bits output from the outer encoder are coupled via a respective constrained interleaver to an input of the inner encoder, and each respective constrained interleaver is configured to implement a respective pseudo-random permutation function to produce a respective permuted output sequence of interleaved bits, wherein each respective pseudo-random permutation function is configured to ensure that at least one common constraint is satisfied, wherein the at least one common constraint limits each respective pseudo-random permutation function by restricting the number of ways output bits may be placed in each respective permuted output sequence of interleaved bits.
 2. The communications apparatus of claim 1, m=ρ₂r₀ρ₁n, where r₁, ρ₁, and n are positive integers, and each multi-bit word corresponds to an r₁ρ₁n-bit word, whereby there are ρ₂ number of r₁ρ₁n-bit words per row.
 3. The communications apparatus of claim 1, wherein any particular multi-bit word corresponds to a multi-bit codeword associated with a corresponding independent encoder that generated the particular multi-bit codeword.
 4. The communications apparatus of claim 1, wherein ρ₂>1.
 5. The communications apparatus of claim 1, wherein ρ₂=1.
 6. The communications apparatus of claim 1, wherein the serially loaded into the r₂×m array of bits along rows corresponds to a row-major order.
 7. The communications apparatus of claim 1, wherein the reading bits out of the r₂×m array along r₂-bit columns corresponds to a column-major order.
 8. The communications apparatus of claim 1, wherein the reading bits out of the r₂×m array along r₂-bit columns is performed from top to bottom in each r₂-bit column during the reading of the bits out of the r₂×m array.
 9. The communications apparatus of claim 1, wherein each of the permutation function, the i^(th) pseudo-random row permutation function, for i=1, 2, . . . r₂, and the j^(th) pseudo-random column permutation function, for j=1, 2, . . . m, corresponds to a respective pseudo randomization that is predetermined prior to a runtime, wherein the runtime corresponds to a time when the constrained interleaver performs the rearrangement of the ordering in accordance with the permutation function.
 10. The communications apparatus of claim 9, further comprising: at least one vector of pointers configured for use, at the runtime, to facilitate the rearrangement of the ordering in accordance with the permutation function.
 11. The communications apparatus of claim 10, wherein the at least one vector of pointers is used to facilitate table lookup operations.
 12. The communications apparatus of claim 1, wherein each respective one of the plurality of the r₂ ρ₂ number of independent encoders is configured to encode a respective sequence of input bits in accordance with a respective code.
 13. The communications apparatus of claim 12, wherein each respective sequence of input bits corresponds to a respective independent sequence of message bits.
 14. The communications apparatus of claim 12, wherein each respective code corresponds to a single common code.
 15. The communications apparatus of claim 13, wherein the single common code corresponds to a serially concatenated code.
 16. The communications apparatus of claim 12, further comprising the r₂ρ₂ number of independent encoders.
 17. The communications apparatus of claim 16, further comprising: an inner encoder configured to receive as input the permuted sequence of output bits and to encode the permuted sequence of output bits in accordance with an inner code.
 18. The communications apparatus of claim 17, further comprising: a signal mapper configured to map an encoded bit sequence to a transmission signal, wherein the signal mapper is a member of the group consisting of a stateless signal mapper and a state-based modulator, and the encoded bit sequence includes coding from the plurality r₂ρ₂ number of independent encoders and from the inner encoder.
 19. The communications apparatus of claim 16, wherein at least one of the r₂ρ₂ number of independent encoders includes an outer encoder that is configured to encode in accordance with an outer code, the outer code is a member of the group consisting of a block code, a LDPC (Low Density Parity Check) code, a convolutional code transformed to a block code, and a non-recursive convolutional code, and the outer code has a minimum Hamming distance of d₀; and a state based modulator configured to transform bits from the permuted sequence of output bits into a sequence of transmit symbols.
 20. The communications apparatus of claim 19, wherein the state-based modulator modulates in accordance with a continuous phase modulation (CPM).
 21. The communications apparatus of claim 16, wherein each respective one of the r₂ρ₂ number of independent encoders corresponds to a respective concatenated encoder that is configured to encode in accordance with a first respective component encoder, a second respective component encoder, and respective constrained interleaver.
 22. The communications apparatus of claim 21, wherein at least one respective pseudo-random permutation function is different from at least one other respective pseudo-random permutation function.
 23. The communications apparatus of claim 21, wherein: the N=r2m input bits comprise N/L number of coded symbols, where N/L and L are positive integers, and each coded symbol comprises L coded bits; and wherein the constrained interleaver is configured so that each i′h pseudo-random row permutation function pseudo-randomly rearranges an ordering of the bits in the i′h row by rearranging the order of L-bit coded symbols on the i′h row, and each j′ pseudo-random column permutation function pseudo-randomly rearranges an ordering of the bits in the j h column by rearranging the order of L-bit coded symbols on the j′t column.
 24. The communications apparatus of claim 23, further comprising: a second inner encoder configured to receive as input the permuted sequence of output bits and to encode the permuted sequence of output bits in accordance with an inner code; wherein the inner code corresponds to a non-binary code.
 25. The communications apparatus of claim 23, further comprising: a second inner encoder configured to receive as input the permuted sequence of output bits and to encode the permuted sequence of output bits in accordance with an inner code.
 26. The communications apparatus of claim 25, wherein the second inner code is a member of the group consisting of a block code, a LDPC (Low Density Parity Check) code, a convolutional code transformed to a block code, and a non-recursive convolutional code.
 27. The communications apparatus of claim 26, wherein: at least one of the r₂ρ₂ number of independent encoders is configured to encode at least partially in accordance with an LDPC code.
 28. The communications apparatus of claim 16, further comprising: a BICM (Bit-Interleaved Coded Modulation) mapper configured to transform a set of input bits onto a respective BICM symbol; wherein the reading bits out of the r₂×m array along the r₂-bit columns to form the permuted sequence of output bits further includes coupling a plurality of subsets of the permuted sequence of output bits as a sequence of sets of input bits to the BICM mapper, and in response thereto, the BICM mapper is configured to produce a sequence of BICM symbols.
 29. The communications apparatus of claim 28, wherein the respective BICM symbol corresponds to an r₂-bit BICM symbol.
 30. The communications apparatus of claim 29, whereby at least d_(o) number of the r₂-bit BICM symbols differ between any two different transmitted sequences of the r₂-bit BICM transmission symbols.
 31. The communications apparatus of claim 29, wherein each BICM symbol is drawn from a 2^(r) ² -ary constellation, and a mapping policy performed by the BICM mapper of the r₂-bit columns is optimized to take into account the permutation function of the constrained interleaver and at least one property of the outer code in order to minimize a measure of bit error probability.
 32. The communications apparatus of claim 28, wherein at least one of the r₂ρ₂ number of independent encoders encodes at least partially in accordance with an LDPC (Low Density Parity Check) code.
 33. The communications apparatus of claim 28, wherein the at least one of the r₂ρ₂ number of independent encoders encodes using the LDPC code as an outer code that is coupled via a constrained interleaver to an inner encoder to form a serial concatenation encoder.
 34. The communications apparatus of claim 28, wherein at least one of the r₂ρ₂ number of independent encoders encodes at least partially in accordance with a member of the group of non-binary codes consisting of a BCH code and a Reed-Solomon (RS) code.
 35. The communications apparatus of claim 28, wherein communications apparatus is further configured to use the sequence of BICM symbols to produce a transmission signal, and the transmission signal is a member of the group consisting of a wireless communication signal and an optical communication signal.
 36. The communications apparatus of claim 35, wherein the transmission signal is a member of the group consisting of a phase-shift keyed (PSK) signal, a quadrature amplitude modulated (QAM) signal, and an orthogonal frequency division multiplexed (OFDM) signal, a multidimensional trellis coded modulation signal, and a multidimensional coded modulation signal.
 37. The communications apparatus of claim 35, wherein the communications apparatus is a member of the group consisting of a headend system configured to communicate with a plurality of user devices, a user device configured to communicate with the headend system, a peer-to-peer communication device, an optical communications device, an optical communications device configured to support a backbone link, an optical communications device configured to support backbone Internet traffic, a cellular communications headend system, a cellular mobile communications device, and a mobile user data enabled communications device.
 38. The communications apparatus of claim 1, further comprising: a signal conditioning unit coupled to receive a received signal and configured to produce therefrom a vector of bit metrics, wherein the received signal is a received version of a transmitted signal is equivalent to one formed using the constrained interleaver to permute the order of a group of N=r₂m outer-encoded bits produced from r₂ρ₂ independent outer encoders to produce a transmitted permuted sequence of output bits; an iterative decoder configured to perform a plurality of iterations of iterative decoding, wherein the iterative decoder comprises: a plurality of deinterleavers that, during each iteration of the iterative decoding, each respectively perform a deinterleaving operation to deinterleave a respective bit sequence that are respective members of the group consisting of bit metrics sequences and extrinsic information sequences, and the deinterleaving operation involves an inverse permutation function that corresponds to an inverse operation of the permutation function; a plurality of decoders that each perform a respective decoding operation related to decoding a respective component code one of an outer code; wherein the iterative decoding comprises passing, via the constrained interleaver and the plurality of deinterleavers, each respective bit sequence for processing by a respective one of the plurality of decoders.
 39. The communications apparatus of claim 38, wherein each respective component code corresponds to a member of the group consisting of an inner code and an outer code.
 40. The communications apparatus of claim 38, wherein each of the plurality of decoders configured to perform a soft decoding operation.
 41. A communications apparatus comprising: a constrained interleaver configured to receive a sequence of N=r₂ρ₂r₁ρ₁n encoded bits that correspond to r₂ρ₂ number of r₁ρ₁n-bit concatenated codewords, and to rearrange the order of the sequence of N=r₂ρ₂r₁ρ₁n encoded bits in accordance with a permutation function in order to produce a constrained-interleaved sequence of encoded bits, wherein N, r₂, ρ₂, ?_(I), ρ_(i), and n are positive integers, wherein the permutation function is configured to pseudo-randomize the order of the sequence of N=r₂ρ₂r₁ρ₁n encoded bits to produce a constrained-interleaved sequence of encoded bits, wherein the permutation function is constrained to ensure that each coded bit of each r₁ρ₁n-bit concatenated codeword is encoded into different respective codewords of an inner code; a set of r₂ρ₂ number of respective concatenated encoders, wherein each respective concatenated encoder is configured to transform a respective sequence of input bits to a respective sub-sequence of r₁ρ₁n number of encoded bits in accordance with a respective concatenated code, wherein each respective sub-sequence of r₁ρ₁n number of encoded bits corresponds to a respective one of the concatenated codewords; an inner encoder configured to transform the constrained-interleaved sequence of outer-encoded bits to a sequence of inner-encoded bits, wherein the sequence of inner-encoded bits is further encoded in accordance with an inner code, and the inner code includes the inner codewords, and each inner codeword is a member of the group consisting of a block codeword, a LDPC codeword, a convolutional code transformed to a block code codeword, and a non-recursive convolutional code codeword; a signal mapper configured to map an encoded bit sequence to a transmission signal, wherein the signal mapper is a member of the group consisting of a stateless signal mapper and a state-based modulator, and the encoded bit sequence includes coding from the set of r₂ρ₂ number of respective concatenated encoders and from the inner encoder; whereby the sequence of inner-encoded bits constitutes a serially-concatenated sequence of bits that incorporates coding from both the inner code and the outer code in accordance with a serially-concatenated code that has a minimum Hamming distance of d_(sc); wherein each respective concatenated code has a minimum Hamming distance of d_(sc1) and the inner code has a minimum Hamming distance of d_(i); wherein the permutation function implemented by the constrained interleaver is constrained to implement a constraint in order to enforce d_(sc)=d_(sc1)d_(i).
 42. The communications apparatus of claim 39, wherein each respective concatenated code is a serially concatenated code.
 43. The communications apparatus of claim 39, wherein each serially concatenated code is a serial concatenation of an outer code that has a minimum Hamming distance of d_(o) and a second inner code that has a minimum Hamming distance of d_(i1), wherein coded bits output from the outer encoder are configured to be coupled via a respective constrained interleaver to an input of a respective second inner encoder to thereby enforce d_(i1)<d_(sc1)=d_(o)d_(i1).
 44. The communications apparatus of claim 41, wherein the outer code encodes using non-binary code symbols.
 45. The communications apparatus of claim 41, wherein the outer code is a non-binary code that is a member of the group consisting of a BCH code and a Reed-Solomon (RS) code.
 46. The communications apparatus of claim 41, wherein the inner code and the second code encode in accordance with a single common code and thus d_(i1)=d_(i).
 47. The communications apparatus of claim 39, wherein the inner code is a systematic code, and each coded bit of each respective concatenated codeword is encoded into different respective codewords of the inner code by placing each coded bit of each respective codeword of the outer code into different respective codewords of the inner code.
 48. The communications apparatus of claim 39, wherein the convolutional code transformed to a block code includes one or more termination bits.
 49. The communications apparatus of claim 46, wherein the one or more termination bits comprises a complete set of termination bits, whereby the convolutional code transformed to a block code is an equivalent block code where the convolutional code is terminated.
 50. A method for use in a communications apparatus that includes a constrained interleaver operative to receive a sequence of N=r₂m input bits that corresponds to r₂ρ₂ number of multibit codewords that were generated from r₂ρ₂ number of independent encoders, and the constrained interleaver is further configured to rearrange an ordering of the sequence of N=r₂m input bits in accordance with a permutation function in order to produce a permuted sequence of output bits, wherein N, r₂, ρ₂, and m are positive integers, the method comprising: rearranging the sequence of N=r₂m input bits in accordance with the permutation function, wherein the permutation function is equivalent to performing the following operations: loading a r₂×m array of bits, wherein the r₂×m array has r rows and m columns, and the bits are serially loaded into the r₂×m array of bits along rows, applying an i^(th) pseudo-random row permutation function to each row i, for i=1, 2, . . . r2, wherein the i^(th) pseudo-random row permutation function pseudo-randomly rearranges an ordering of the bits in the i^(th) row, applying a j^(th) pseudo-random column permutation function to each r₂-bit column j, for j=1,2, . . . m, wherein the j^(th) pseudo-random column permutation function pseudo-randomly rearranges an ordering of the bits in the j^(th) r₂-bit column, and reading bits out of the r₂×m array along r₂-bit columns to form the permuted sequence of output bits; wherein each of the permutation function, the i^(th) pseudo-random row permutation function, for i=1, 2, . . . q, and the i^(th) pseudo-random column permutation function, for i=1,2, . . . m, corresponds to a respective pseudo randomization that is predetermined prior to a runtime, wherein the runtime corresponds to a time when the method is operative.
 51. The method of claim 50, m=ρ₂r₁₂ρ₁n, where r₁ρ₁, and n are positive integers, and each multi-bit word corresponds to an r₁ρ₁n-bit word, whereby there are ρ₂ number of r₁ρ₁n-bit words per row.
 52. The method of claim 50, wherein any particular multi-bit word corresponds to a multi-bit codeword associated with a corresponding independent encoder that generated the particular multi-bit codeword.
 53. The method of claim 50, further comprising: using at least one vector of pointers to facilitate the rearranging.
 54. The method of claim 53, wherein using comprises table lookup operations.
 55. The method of claim 50, further comprising; encoding, in accordance with the plurality of independent encoders, a respective plurality of sequences of input bits to generate the sequence of N=r₂m input bits.
 56. The method of claim 55, further comprising: encoding, in accordance with an inner encoder, the permuted sequence of output bits.
 57. The method of claim 55, further comprising: mapping, in accordance with a Bit-Interleaved Coded Modulation (BICM) mapping function, a sequence of respective groups of bits from the permuted sequence of output bits to a respective sequence of BICM transmit symbols.
 58. The method of claim 56, further comprising: mapping a sequence of encoded bits to a transmission signal, wherein the mapping is a member of the group consisting of a stateless signal mapping and a state-based modulation, the encoded bit sequence includes coding from the plurality of independent encoders and from the inner encoder, and the transmission signal is a member of the group consisting of an optical signal and a wireless signal.
 59. The method of claim 50, further comprising: receiving a received signal and producing therefrom a vector of bit metrics, wherein the received signal is a received version of a transmitted signal that was serially-concatenated encoded by a serially-concatenated encoder that coupled an outer encoded bit stream via a second constrained interleaver to an inner encoder, wherein the second constrained interleaver implements the permutation function; and iteratively decoding, using a plurality of component decoders and the constrained interleaver, at least a plurality of deinterleaved a respective bit sequences that are respective members of the group consisting of bit metrics sequences and extrinsic information sequences, and performing the deinterleaving operations that involve an inverse permutation function that corresponds to an inverse operation of the permutation function. 